Electrónica de Potencia

ELECTRÓNICA DE POTENCIA/ ELECTRÓNICA INDUSTRIAL 1 Y LABORATORIO 2547820/2507385 Por Pedro Vicente Leal Pérez Ingeniero Electrónco U de A Especialista en Telemática U de A Profesor TC Universidad de Antioquia Facultad de Ingeniería Departamento de Electrónica y Telecomunicaciones

Referencias bibliográficas: 1. Rashid, Muhammad, H. Electrónica de potencia: circuitos, dispositivos y aplicaciones. Pearson Educación, 2004 - 908 p.

2. Hart, Daniel, W. Electrónica de Potencia. Pearson Educación, S. A. Madrid. 2001-472 p 3. Gunasekaran, MK. Circuits for Analog System Design. Center for Electronic Design and Tecnologic. Indian Institute of Science.

Bangalore. 2012-2019 4. Mohan Ned y otros. Electrónica de Potencia: Convertidores, aplicaciones y diseño.Tercera Edición.McGawhill/interamericana editores, s.a. de c.v. 2009

N. Mohan, T. M. Undeland, W.P Robbins, ―Power Electronics, Converters, Applications& Design,‖ Wiley India Pvt. Ltd. 2. M. H. Rashid, ―Hand book of Power Electronics” , Academic Press,2001.

3. Daniel.W.Hart, "Power Electronics", Mc GrawHill Publications 2010.

4. Joseph Vithayathil, ―Power Electronics‖, Tata McGraw Hill.

5. P.S Bhimbra, "Power Electronics",Khanna Publishers.

6. Simon Ang, Alejandro Oliva, "Power-Switching Converters" Taylor and Francis group 7. R W Erickson and D Maksimovic, ―Fundamental of Power Electronics‖ Springer, 2nd Edition.

Otras referencias 1. P. T. Krein, ―Elements of Power Electronics‖, Oxford University Press.

2. L.Umanad, "Power Electronics: Essentials & Applications," Wiley.

3. IEEE Transaction journals, IECON, APEC and other power electronic related Conference Proceedings etc.

Referencias de Internet: 1. 3. Gunasekaran, MK. Circuits for Analog System Design. Center for Electronic Design and Technology. Indian Institute of Science.

2. Power Supply and Protection. CircuitsCircuits for Analog System Design. Prof Gunasekaran M.K. Center for Electronic Design and Technology. Indian Institute of Science. Bangalore.

3. https://04.phf-site.com/2016/12/m-k-gunasekaran-electronicscircuits.html 4. Power Semiconductor Devices.Version 2 EE IIT, Kharagpur.documents.pub_power-electronics-5584a51b377f7

Oficina: 19-407 Pedro.leal@udea.edu.co Atención a estudiantes: L, M, Mi, J, V, 10-12M ,

L, Mi: 14-18 PM

Atención y asesoría en Tecnologia, Instrumentación, simulación en SPICE y LabView, Montajes y aspectos Teóricos y prácticos del curso: Ingeniero Electrónico Sergio Andrés Díaz Correa 20-129 sergio.diaz@udea.edu.co

PROGRAMA RESUMIDO UNIDAD UNO Definición intuitiva y rigurosa de la Electrónica de Potencia. Fuentes primarias de energía tradicionales y perspectivas futuras. Tipos de conversión de Energía Eléctrica. Primitivas en la Electrónica de Potencia.

UNIDAD DOS Conversión AC/DC con entrada monofásica y polifásica, para carga diversa: R, RL. RC, RLV. Topologías. Tratamiento matemático, tratamiento simulado, modelaje de interruptores, cálculo y diseño computacional con funciones objetivo en SPICE. Ayuda del ambiente LabView para la generación de formas de onda de potencia, simuladas sin y con varición de parámetros. Utilización del programa SPICE en forma intensiva para los efectos citado.

UNIDAD TRES Conversión AC/AC, AC/DC con interruptores no controlados y controlados de estado sólido. Topologías. Encendido-apagado asincrónico y sincrónico de los interruptores. Parámetros, modelos, hoja de datos y regiones de trabajo. Formas de onda y cálculos manuales y computacionales. Diseño Electrónico Asistido por Computador

UNIDAD CUATRO Conversión DC/DC y DC/AC básicas con dispositivos de estado sólido.

Topologías. cálculos manuales y computacionales. Diseño Electrónico Asistido por Computador UNIDAD CINCO Protecciones Electrónicas, , tanto de lazo abierto como de lazo cerrado, para equipos y sistemas de baja energía. Topologías.

Cálculos manuales y computacionales.

UNIDAD SEIS Sistemas de control calibrado y automático para máquinas eléctricas , DC, técnicas de control por fase, técnica PWM, con envío de consignas desde o fuera de plataforma computacional. Drivers y topologías diversas.

DESARROLLO DE UNIDADES UNIDAD UNO: Definición intuitiva y rigurosa de la Electrónica de Potencia. Fuentes primarias de energía tradicionales y perspectivas futuras. Tipos de conversión de Energía Eléctrica. Primitivas en la Electrónica de Potencia.

PE DEFINICIÓN INTUITIVA: Ciencia y tecnología electrónica que se ocupa de aplicar la Electrónica a la solución de los problemas de conversión de la Energía Eléctrica EN FORMA RIGUROSA: 1. “La electrónica de potencia es la aplicación de la electrónica de estado sólido para controlar y convertir una forma de energía eléctrica en otra, como convertir entre CA y CC o cambiar la magnitud y fase del voltaje y la corriente o frecuencia o una combinación de estos” Tomado de: Blaabjerg Blaabjerg, Frede. Control of Power Electronic Converters and Systems. Academic Press. Published Date: 24th January 2018. 192 p 39 2.

2. “La electrónica de potencia implica el estudio de circuitos electrónicos destinados a controlar el flujo de energía eléctrica”.

Estos circuitos manejan un flujo de energía a niveles mucho más altos que los niveles manejados por dispositivos de uso común.

Tomado de: Larry Francis Obando – Technical Specialist – Educational Content Writer

3. Ciencia y tecnología electrónica, principalmente de estado sólido, que se ocupa de la conversión o flujo controlado de energía eléctrica, desde una forma a otra, o modificando alguno de sus parámetros.

4. Power Electronics (IEEE) “Power electronics can be defined as the use of electronic devices to control and convert electric power. Therefore, every application of power electronics involves some aspect of control. Since the mathematical tools used to solve one control problem are often common to other problems, some researchers specialize in formulating control algorithms for a variety of power electronic applications. Therefore, it is useful to review the major areas of power electronics from the control perspective, in order to understand the broadbased connection between modern control theory and power electronics as a whole.” (Home » Power Electronics » Control Theory)

FUENTES PRIMARIAS DE ENERGÍA TRADICIONALES Y PERSPECTIVAS FUTURAS Hidráulica:

hecho en canva

Se acumula agua en un embalse y se aprovecha la conversión de energía potencial por altura, en cinética para mover las aspas de un generador eléctrico

Térmica

hecho en canva

Mediante la combustión del carbón, se hacen mover las aspas de generador eléctrico

Eólica

Imagen de uso libre La fuente primaria es el viento en aerogeneradores eléctricos Sola

r Es la conversión directa de la energía solar en eléctrica por medio de celdas solares

Maremptriz.

Hecho en canva La fuente primaria es la producida por las olas del mar Química.

La energía utilizada son acumuladoresquímicos de donde se libera energía

Gaseomotriz.

La energía utilizada es la combustión de un gas

Atómica.

La energía utilizada es la de operaciones de fisión y fusión

l futuro son las fuentes de energía limpias y renovables como la de la fusión nuclear, tratando de replicar los procesos solares, usando por ejemplo el isótopo 3 del Helio depositado por años en la luna.

PRINCIPALES DISPOSITIVOS DE ELECTRÓNICA DE POTENCIA Principal Power Electronic Devices

SIT power Static Induction Transistor GTO MCT which is a high power, high frequency MOS- COOLMOS gated turn-on and turn-off gate source voltage VGS MCTs, the P-MCT and the N-MCT HEXFET d IR Transistor VMOS Transistor DMOS Transistor TMOS Transistor BIMOS Hibrid Bipolar and MOS GaN Power Semiconductor GaN FET Dynamic SiC-based power semiconductor solutions GTR GTR Power Electronics Giant Transistor DIACS AC Diode LADIAC

Protection . Fuse and circuit breakers; Electromagnetic relays; Solid State Relays; Optical Isolators; Over-voltage protection – MOVs; Over-current protection; Circuit Supplies: Power Management Surge Protection for Power Electronic Systems.

PD. Diodos de potencia. Power Diodes(Uncontrolled device Diode-Clamped GCTs and Voltage Equalization FRRD. Diodo de Recuperación Inversa Rápido. Fast Reverse Recovery Diod, trr is usually less than 1μs, for many less than 100 ns Rectifier diode General purpose diode Giant Transistor—GTR GTR, GTR is also called power BJT, or just BJT BJT. Tansistores bipolares de potencia..Power Bijunction or Power JFET. Transistores FET de juntura de potencia. PowerJunction Field Effect Transistor MOSFET. Transistores de efecto de campo de potencia. Metal Oxide Semiconductor Field Effect Transistor Power MOSFET Power MOSFET transistor

Power

metal-oxide-semiconductor

field

effect

IGBT Transistor Bipolar de potencia de Compuerta Aislada Power Insulated-Gate Bipolar Transistor

Dentro de la familia de Tiristores están los siguientes: SCR. Rectificador Controlado de Silicio. Silicon Controlled Rectifier

DISPOSITIVOS CONTROLADOS TÍPICOS Typical fully-controlled devices

LASCR. SCR activado por Luz- Light Activated SCR GTO. Tiristor de encendido y apagado por Gate. Gate Trigger Off= Gateturn-off thyristor MCT. Tiristor MOSFET Controlado. Mosfet Controlled Thyristor LASCS. Suiche Controlado de Silicio Activado por Luz. Light Activated Silico Static induction transistor —} SIT. Static induction thyristor n Controlled Switch Smart power integrated circuit(Smart power IC, SPIC, Smart switch) PEM power electronics Module Other power electronic devices SIT Static induction transistor SITH MOS controlled thyristor Static induction thyristor — SITh. Tiristor de Inducción Estática. R, Static Induction Thyristor RCT. Tiristor controlado en forma inversa. Reverse Contrlled Thyristor

SDAC. Diodo de Avalancha de cuatro capas. Slicon Diode AC . TRIAC. Triodo AC. AC Triode LATRIAC+. Triac Activado por Luz. Light Activated AC Triode MTO . Tiristor de apagado por MOS. MOS Turn Off Thyristor GTR IGCT. Tiristor de conmutación por Circuito Integrado de Gate .

Integrated Gate Conmutated Thyristor (SITH) Static Induction Thyristor TRIAC – TRIode for Alternating Current DIAC – DIode on AC MOSFET – Metal Oxide Semiconductor Field Effect Transistor IGBT – Insulated Gate Bipolar Transistor SIT – Static Induction Transistor SITH – Static Induction Thyristor IGCT – Integrated Gate Commutated Thyristor ( also known as GCT) LTT (LASCR) – Light Triggered(Activated) Thyristor (MOSFET Metal-Oxide-Semiconductor Field-Effect Transistor de agotamiento

CARACTERÍSTICAS GRÁFICAS Y REGIONES DE TRABAJO DE ALGUNOS DISPOSITIVOS DE POTENCIA Y/O DE AYUDA

DIODO RECTIFICADOR

BJT

Tomado de: Muñoz L. Oscar A.López Z. ustavo A.exto Guía de Electrónica de Potencia. Trabajo de grado, Corporación Universitaria Autónoma de Occidente, Santiago de Cali, 1997

MÁXIMA DISCIPACIÓN DE POTENCIA DE BJT Tomado de: Muñoz L. Oscar A.López Z. ustavo A.exto Guía de Electrónica de Potencia. Trabajo de grado, Corporación Universitaria Autónoma de Occidente, Santiago de Cali, 1997

SCR

Tomado de: Muñoz L. Oscar A.López Z. ustavo A.exto Guía de Electrónica de Potencia. Trabajo de grado, Corporación Universitaria Autónoma de Occidente, Santiago de Cali, 1997

MOSFET CANAL N

IGBT

Tomado de:

Hart, W, Daniel. Introducción a la Electrónica de Potencia. Prentice Hall es un sello editorial autorizado dePpearson Educación, s. a. 2001.

472 p

Tomado de:

Hart, W, Daniel. Introducción a la Electrónica de Potencia. Prentice Hall es un sello editorial autorizado dePpearson Educación, s. a. 2001.

472 p

Tomado de:

Hart, W, Daniel. Introducción a la Electrónica de Potencia. Prentice Hall es un sello editorial autorizado dePpearson Educación, s. a. 2001.

472 p

Tomado de: Hart, W, Daniel. Introducción a la Electrónica de Potencia. Prentice Hall es un sello editorial autorizado dePpearson Educación, s. a. 2001.

472 p

TIPOS DE CONVERSIÓN DE LA ENERGÍA ELÉCTRICA

PRIMITIVAS EN LA ELECTRÓNICA DE POTENCIA. La carga eléctrica, q, es una consecuencia del desequilibrio entre protones y electrones en un pedazo de material u elemento, y sis dicha carga se mueve al aplicar campos eléctricos, magnéticos o electro-magnéticos, eso constituye corriente ,i, eléctrica interna o externa , entonces i=dq/dt, con: q=carga en coulombs t=tiempo i=corriente en A o mA d/d es la derivada

Para que haya corriente se requiere una diferencia de potencial o voltaje entre dos puntos cualesquiera, y esta tensió o voltaje ,v, se define como un cambio de e la energía potencial que experimenta la unidad de carga, y se expresa en forma matemática, así: V=d(Ep)/dq v=voltaje en voltios q en coulombios Ep en julios o ergios

La potencia intantánea de define como la velocidad a la cual se gasta o acumula energía, y se utiliza la p para ello P=v*i=(dEp/dq)dq/dt=dEp/dt Siendo p en watts t= segundos Ep en Julios o ergios

Si parimos de la expresión p=v*i=(dEp/dq)dq/dt=dEp/dt

p(t)*dt=dEp que al integrar entre 0 y t con Ep(o) y Ep(t), obtenemos

Valor medio de la potencia

Valor eficaz de una forma de onda v/t) Tomado de: Hart, W, Daniel. Introducción a la Electrónica de Potencia. Prentice Hall es un sello editorial autorizado dePpearson Educación, s. a. 2001.

472 p

Potencia aparente s

Descomposición de Fourier

Tomado de: Hart, W, Daniel. Introducción a la Electrónica de Potencia. Prentice Hall es un sello editorial autorizado dePpearson Educación, s. a. 2001.

472 p Factor de potencia =FP Rizado de una señal=Valor rms de los armónicos/valor dc

Rizad=RMS de la componente ac/valor dc %rizad=rizado*100% Identidad de Parseval (RMS true)2 =(DC)2 +(RMS de armónicos)2 La Distorsión Armónica Total (THD) es una medida de cuánto se «distorsiona» o se cambia una forma de onda al pasar por un procesdor de señal o de potencia

Tomado de: https://es.wikipedia.org/wiki/Distorsi%C3%B3n_arm%C3%B3nica

DESARROLLO DE UNIDAD DOS PE Conversión AC/DC no controlada y controlada con entrada monofásica y polifásica, para carga diversa: R, RL. RC, RLV. Topologías. Tratamiento matemático, tratamiento simulado, modelaje de interruptores, cálculo y diseño computacional con funciones objetivo en SPICE. Ayuda del ambiente LabView para la generación de formas de onda de potencia, simuladas sin y con varición de parámetros. Utilización del programa SPICE en forma intensiva para los efectos citado.

Conversión AC/DC, no controlada y controlada, con entrada monofásica y polifásica, para carga diversa: R, RL. RC, RLV. Topologías.

Factor de potencia =FP Rizado de una señal=Valor rms de los armónicos/valor dc Rizad=RMS de la componente ac/valor dc %rizad=rizado*100% Identidad de Parseval (RMS true)2 =(DC)2 +(RMS de armónicos)2 La Distorsión Armónica Total (THD) es una medida de cuánto se «distorsiona» o se cambia una forma de onda al pasar por un procesdor de señal o de potencia

V(t) es la entrada, para el caso senoidal monofásica

V(t)=Vm*sin(wt) W=2*π*f Con f=frecuencia en Hz T=tiempo en segundos Vm= amplitud o valor máximo de la excitación

Onda generada con el código SPICE HOLA V1 1 0 SIN(0 10 60 0 0 0) R1 1 0 1K .PROBE .TRAN 41u 16.66m 0 41u .end DISPOSITIVOS SEMICONDUCTORES DE POTENCIA Imágenes tomadas con fines académicos de: 1. The Basics of Power Semiconductor Devices: Structures, Symbols, and Operations June 13, 2015 by Editorial Team 2. https://tallerelectronica.com/transistor-jfet/

DIODE= PN Power Semiconductor Junction

SCR=Silicon Controlled Rectifier= THYRISTOR

TRIODE= AC Triode=THYRISTOR

BJT= Bi-junction Transistor= Bipolar Transistor

JFET= Junction Field Effect Transistor

MOSFET= Metal Oxide Semiconductor Field Effect Transisitor

LASCR= Light Activated SCR

LATRIAC= Light Activated TRIAC

GTO= Gate Turn-off Thyristor

IGCT=Integrated Gate-Commutated Thyristor

IGBT=Insulated Gate Bipolar Transistor

SMART POWER MOSFET= Inteligent Mosfet

CARACTERÍSTICA DEL SCR

Rectificador no controlado monofásico carga R

Código SPICE texto Hola V1 1 0 sin(0 156 60 0 0 0) D1 1 2 d1N4004 .lib diode.lib R1 2 0 10k .tran 41u 16.66m 0 41u .probe

.end

Gráfico de salida v(2) salida no controlada

Rectificador controlado monofásico controla con carga R

Rectificador controlado monofásico carga R Código SPICE Hola v1 1 0 sin(0 156 60 0 0 0) r1 3 0 1k x1 1 2 3 2n1599 .lib THYRISTR.lib v2 2 3 pulse(0 5 {tdisp}1n 1n 10u 16.66m) .param tdisp=1m .step param tdisp 1.38m 8m 1m .tran 41u 16.66m 0 41u .probe .end

Salida controladan en R1 para disparo en 30°

Parámetros del tiristor utilizado .subckt 2N1599-X

anode gate cathode

* "Typical" parameters X1 anode gate cathode Scr params: + Vdrm=400v Vrrm=400v + dVdt=1e9 Igt=2ma

Ih=5ma

Vgt=.7v

Vtm=1.1v Ton=0.8u

+ Idrm=10u * 90-5-18 .ends *$

Motorola

DL137, Rev 2, 3/89

Itm=1

Toff=10u

Rectificador no controlado con carga R, L serie

Código SPICE rexto Hola V1 1 0 sin(0 156 60 0 0 0) D1 1 2 d1N4004 .lib diode.lib R1 2 3 100 L1 3 0 50mh ic=0 .tran 41u 16.66m 0 41u .probe .end

Voltaje de entrada y de salida con oscilaciones

Gráfico de corriente en la carga, con conducción superior a 180°

Rectificador monofásico carga R, L serie con diodo de giro libre Para reducir las oscilaciones

Código SPICE

Hola V1 1 0 sin(0 156 60 0 0 0) D1 1 2 d1N4004 D2 0 2 d1N4004 .lib diode.lib R1 2 3 100 L1 3 0 50mh ic=0 .tran 41u 16.66m 0 41u .probe .end

Salida sin oscilaciones

La corriente en R!

La corriente en el diodo D1

Rectificador monofásico no controlad de onda completa, carga R, L serie, con condición inicial en la bobina D1, D2, D3, D4 D1N4004

Ódigo SPICE texto

hola v1 1 2 sin(0 25.4 60 0 0 0 d1 1 3 d1n4004 d2 2 3 d1n4004 d3 0 1 d1n4004 d4 0 2 d1n4004 .lib diode.lib r1 3 4 100 l1 4 0 100mh ic=44.76ma .probe .tran 41u 16.66m 0 41u uic .end

Salida en nodo 3

Corriente en la resistencia de carga

Rectificador mpnofásico controlado de onda completa carga R,L serie, conducción discontinua , o sea ic=0 para la corriente en la bobina

Código SPICE texto

Hola V1 1 2 sin(0 156 60 0 0 0) D1 0 1 D1N4004 D2 0 2 D1N4004 D3 0 3 D1N4004 .lib diode.lib X1 1 5 3 2N1599 X2 2 6 3 2N1599 .lib THYRISTR.lib V2 5 3 pulse( 0 5 4.16m 1n 1n 20u 16.66m) V3 6 3 pulse(0 5 12.49m 1n 1n 20u 16.66m) R1 3 4 100 L1 4 0 100mh .tran 41u 16.66m 0 41u .probe .end

Salida con el diodo de giro libre

Retificación monofásica controlada onda completa, conducción continua, ángulo de disparo 15° y simulación de 50m

Código SPICE

Hola V1 1 2 sin(0 156 60 0 0 0) D1 0 1 D1N4004 D2 0 2 D1N4004 ;D3 0 3 D1N4004 .lib diode.lib X1 1 5 3 2N1599 X2 2 6 3 2N1599 .lib THYRISTR.lib Vg1 5 3 pulse( 0 5 0.69m 1n 1n 20u 16.66m) Vg2 6 3 pulse(0 5 9.02m 1n 1n 20u 16.66m) R1 3 4 100 L1 4 0 100mh .tran 100u 50m 0 100u .probe .end

Salida de corriente

Voltaje de salida

Rectificador controlado trifásico de media onda carga R, conducción continua con ángulo de disparo entre 0° y 30° respecto a los puntos de conmutación natural

Código SPICE hola v1 1 0 sin(0 156 60 0 0 0) v2 2 0 sin(0 156 60 0 0 -120) v3 3 0 sin(0 156 60 0 0 -240) x1 1 4 7 2N1599 x2 2 5 7 2N1599 x3 3 6 7 2N1599 .lib THYRISTR.lib Vg1 4 7 pulse(0 5 2.08m 1n 1n 10u 16.66m) Vg2 5 7 pulse(0 5 7.63m 1n 1n 10u 16.66m) Vg3 6 7 pulse(0 5 13.18m 1n 1n 10u 16.66m) R1 7 0 220 .probe .tran 41u 16.66m 0 41u .end

Salida incluyendo v3 al principio

Temas relacionados Valor medio y Valor eficaz El optoscr Rectificador monofásico de media onda controlado Rectificador monofásico de onda completa semicontrolado Rectificador monofásico de onda completa controlado Rectificador trifásico de media onda controlado Rectificador trifásico de onda completa semicontrolado Rectificador trifásico de onda completa controlado El optotriac Controlador ac monofásico de onda completa Troceador clase A

hola v1 1 0 sin(0 56 60 0 0 0) v2 2 0 sin(0 56 60 0 0 -120) v3 3 0 sin(0 56 60 0 0 -240) d1 1 4 d1n4004 d2 2 4 d1n4004 d3 3 4 d1n4004 d4 5 1 d1n4004 d5 5 2 d1n4004 d6 5 3 d1n4004 .lib diode.lib

R1 4 5 1k .probe .tran 41u 16.66m 0 41u .end

https://wilaebaelectronica.blogspot.com/2017/05/rectificador-trifasicode-onda-completa-semicontrolado.html

RAMPA INVERTIDA PARA EL NIVEL Hola V1 1 0 pulse(5 0 0 4 1n 1n 4) R1 1 0 1k .tran 16m 8 0 16m .probe .end

CONTROLADO DE MEDIA ONDA

ONDA COMPLETA CONTROLADO

https://wilaebaelectronica.blogspot.com/2017/05/rectificador-trifasicode-onda-completa-semicontrolado.html

REEMPLAZO DE THYRISTORES POR DIODOS

Índice de contenido Por Wilaeba electronica julio 09, 2020

Amplificador Operacional Aplicaciones clásicas 01. Amplificador Seguidor de Voltaje 02. Amplificador Comparador Temporizador con amplificador operacional 03. Amplificador Inversor 04. Amplificador No inversor 05. Amplificador Sumador inversor 06. Amplificador Sumador no inversor 07. Amplificador Sumador Restador Amplificador restador 08. Amplificador integrador 09. Amplificador derivador Aplicaciones no clásicas 01. Conversor digital analogico por suma ponderada 02. Conversor digital analogico R-2R 03. Conversor analógico digital flash (Paralelo) 04. Conversor analógico digital por rampa digital

05. Conversor analógico digital por seguimiento 06. Conversor analógico digital por aproximaciones sucesivas 07. Disparador Schmitt no inversor 08. Disparador Schmitt inversor 09. Amplificador operacional con fuente única 10. Rectificador de precisión de media onda 11. Rectificador de precisión de onda completa 12. Amplificador desfasador 13. Amplificador de instrumentación Compensación de unión fría Osciladores 01. Oscilador por corrimiento de fase en adelanto con Opamp 02. Oscilador por corrimiento de fase en atraso con Opamp 03. Oscilador en cuadratura 04. Oscilador puente de Wien 05. Oscilador de relajación con amplificador operacional PWM variable con amplificador operacional 06. Generador de señal triangular y señal cuadrada Generador de señal diente de sierra y PWM variable 07. Oscilador Colpitts con Amplificador operacional 08. Oscilador Hartley con Amplificador operacional Controles de temperatura Control on off de temperatura sin histéresis v1.0 Control on off de temperatura sin histéresis v2.0

Control on off de temperatura con histéresis v1.0 Control on off de temperatura con histéresis v2.0

Digitales Basico Compuertas logicas 01. Compuerta lógica Not 02. Compuerta lógica And 03. Compuerta lógica Or 04. Compuerta lógica Xor 05. Compuerta lógica Nand 06. Compuerta lógica Nor 07. Compuerta lógica Xnor Algebra de boole Lógica combinacional 01. Sumador Binario Restador de dos números de 4 bits Sumador restador de dos números de 4 bits Sumador de dos números bcd Restador de dos números bcd Sumador restador de dos números bcd 02. Multiplicador de dos números de 2 bits 03. Multiplicador de dos números de 3 bits 04. Multiplicador de dos números de 4 bits 05. Hola en display 7 segmentos

06. Decodificador bcd a 7 segmentos 07. Decodificador hexadecimal a 7 segmentos 08. Decodificador Binario a bcd de 4 bits 09. Decodificador Binario a bcd de 5 bits 10. Decodificador Binario a bcd de 6 bits 11. Decodificador Binario a bcd de 7 bits 12. Decodificador Binario a bcd de 8 bits 13. Comparador de dos números de 1 bit Comparador de dos números de 2 bits Comparador de dos números de 3 bits Comparador de dos números de 4 bits CI 555 (Aplicaciones típicas) 01. Generador de señal de reloj (CLK) con el CI 555 02. Frecuencia variable con el CI 555 03. PWM variable con el CI 555 04. Temporizador con el CI 555 05. Temporizador cíclico con el CI 555 Contadores 02. Contador asíncrono de década ascendente 03. Contador asíncrono de 4 bits ascendente 04. Contador asíncrono de 4 bits descendente 05. Contador síncrono de década ascendente 06. Contador síncrono de década descendente 07. Contador síncrono de década ascendente descendente 08. Contador síncrono de 4 bits ascendente

09. Contador síncrono de 4 bits descendente 10. Contador síncrono de 4 bits ascendente descendente Contadores en cascada 01. Contador 1 12 ascendente parte 1/2 02. Contador 1 12 ascendente parte 2/2 03. Contador 0 23 ascendente parte 1/2 04. Contador 0 23 ascendente parte 2/2 05. Contador 0 59 ascendente parte 1/2 06. Contador 0 59 ascendente parte 2/2 07. Contador 0 59 descendente 08. Contador 0 59 ascendente descendente 09. Contador 0 99 ascendente parte 1/2 10. Contador 0 99 ascendente parte 2/2 11. Contador 0 99 descendente 12. Contador 0 99 ascendente descendente

Filtros analogicos 1. Filtros Pasa altos 1. Filtro pasa altos Pasivo 1er Orden RC 2. Filtro pasa altos Pasivo 1er Orden RL 3. Filtro pasa altos Pasivo 2do Orden RC 4. Filtro pasa altos Pasivo 2do Orden RLC 5. Filtro pasa altos Activo 1er Orden RC 6. Filtro pasa altos Activo 2do Orden Sallen Key

2. Filtros Pasa bajos 1. Filtro pasa bajos Pasivo 1er Orden RC 2. Filtro pasa bajos Pasivo 1er Orden RL 3. Filtro pasa bajos Pasivo 2do orden RC 4. Filtro pasa bajos Pasivo 2do Orden RLC 5. Filtro pasa bajos Activo 1er Orden RC 6. Filtro pasa bajos Activo 2do Orden Sallen Key 7. Filtro pasa bajos Activo 2do Orden MFB 3. Filtros Pasa banda 1. Filtro pasa banda Pasivo 2do Orden RC 2. Filtro pasa banda Pasivo 2do Orden RLC 3. Filtro pasa banda Activo 2do Orden RC 4. Filtro pasa banda Activo 2do Orden Sallen Key 5. Filtro pasa banda Activo 2do Orden MFB 6. Filtro pasa banda Activo 2do Orden con Amplificador dual 4. Filtros Rechaza Banda 1. Filtro Rechaza Banda Pasivo RC Twin-T 2. Filtro Rechaza Banda Pasivo 2do Orden RLC 3. Filtro Rechaza Banda Activo 2do Orden RC 4. Filtro Rechaza Banda Activo RC Twin-T (Circuito A) 5. Filtro Rechaza Banda Activo RC Twin-T (Circuito B) 6. Filtro Rechaza Banda Activo 2do Orden Sallen Key (Circuito A) 7. Filtro Rechaza Banda Activo 2do Orden Sallen Key (Circuito B) 8. Filtro Rechaza Banda Activo 2do Orden MFB 5. Aproximaciones

1. Aproximación Butterworth 2. Aproximación Chebyshev 6. Filtros de Orden superior 1. Filtros Pasivos de orden superior Pasa bajos 2. Filtros Pasivos de orden superior Pasa altos 3. Filtros Pasivos de orden superior Pasa banda 4. Filtros Pasivos de orden superior Rechaza banda 5. Filtros Activos de Orden superior Pasa bajos 6. Filtros Activos de Orden superior Pasa altos 7. Filtros Activos de Orden superior Pasa banda 8. Filtros Activos de Orden superior Rechaza banda 7. Filtros Activos universales (Multifunción) 1. Filtro Bicuadrático KHN Inversor 2. Filtro Bicuadrático KHN No inversor 3. Filtro Bicuadrático Tow - Thomas

Filtros básicos de audio 1. Filtros High Shelf 1. Filtro High Shelf de 1er orden RC tipo I 2. Filtro High Shelf de 1er orden RC tipo II 2. Filtros Low Shelf Filtro Low Shelf de 1er orden RC 3. Filtros Peak-Notch 1. Filtro Peak-Notch con Red RLC

2. Filtro Peak-Notch con Red C-Girador 3. Filtro Peak-Notch con BPF Sallen Key 4. Control de tonos 1. Control de tonos I (tipo paralelo) 2. Control de tonos II (tipo paralelo)

Transistor BJT 1. Polarizacion transistor bjt 1. Polarización fija bjt 2. Polarización estabilizado en emisor 3. Polarización por divisor de voltaje bjt 4. Polarización por realimentación de colector 5. Polarización por realimentación de colector y de emisor 6. Transistor BJT como conmutador 2. Análisis ac transistor bjt 1. Configuración emisor común bjt 1. Emisor común con polarización fija bjt 2. Emisor común con polarización estabilizado en emisor 3. Emisor común con polarización por divisor de voltaje bjt 4. Emisor común con polarización por realimentación de colector 5. Emisor común con polarización por realimentación de colector y de emisor 2. Configuración base común bjt 1. Base común con polarización estabilizado en emisor

2. Base común con polarización por divisor de voltaje 3. Base común con polarización por realimentación de colector y de emisor 3. Configuración colector común bjt 1. Colector común con polarización estabilizado en emisor 2. Colector común con polarización por divisor de voltaje 3. Colector común con polarización por realimentación de colector y de emisor 4. Efecto de la resistencia de fuente RS 5. Efecto de la resistencia de carga RL 6. Efecto de las resistencias de fuente y de carga (RS y RL) 3. Osciladores con transistor bjt 2. Oscilador por corrimiento de fase con Transistor bjt

Transistor JFET 1. Polarizacion transistor JFET 1. Polarización fija JFET 2. Autopolarización JFET 3. Polarización por divisor de voltaje JFET 2. Análisis ac transistor JFET 1. Fuente común con polarización fija JFET 2. Fuente común con autopolarización JFET 3. Fuente común con polarización por divisor de voltaje JFET 4. Compuerta común con autopolarización JFET

5. Compuerta común con polarización por divisor de voltaje JFET 6. Drenaje común con autopolarización JFET 7. Drenaje común con polarización por divisor de voltaje JFET

Control analogico 1. Diagrama de bloques 1. Diagrama de bloques - Solución por ecuaciones algebraicas 2. Diagrama de bloques - Solución por álgebra de bloques 3. Diagrama de bloques - Solución por fórmula de ganancia de Mason 2. Sistemas de segundo Orden 1. Sistema de segundo orden subamortiguado (0 < ζ < 1) 2. Sistema de segundo orden sobreamortiguado (ζ ≥ 1) 3. Planta de segundo orden con red RC 4. Planta de segundo orden con red RLC 5. Planta de segundo orden con Amplificador operacional ( Filtro Sallen Key ) 6. Planta de segundo orden con Amplificador operacional ( Filtro MFB ) 3. Compensadores de fase 1. Circuitos compensadores de fase 1. Circuito compensador de adelanto de fase no inversor 2. Circuito compensador de adelanto de fase inversor 3. Circuito compensador de atraso de fase no inversor

4. Circuito compensador de atraso de fase inversor 5. Circuito compensador de adelanto - atraso de fase no inversor 6. Circuito compensador de adelanto - atraso de fase inversor 2. Diseño de compensadores de fase 1. Diseño de compensador de adelanto por el método de lugar geométrico de las raíces Ejemplo 1. Diseño de compensador de adelanto por el método de lugar geométrico de las raíces Ejemplo 2. Diseño de compensador de adelanto por el método de lugar geométrico de las raíces Ejemplo 3. Diseño de compensador de adelanto por el método de lugar geométrico de las raíces Ejemplo 4. Diseño de compensador de adelanto por el método de lugar geométrico de las raíces Ejemplo 5. Diseño de compensador de adelanto por el método de lugar geométrico de las raíces 2. Diseño de compensador de atraso por el método de lugar geométrico de las raíces Ejemplo 1. Diseño de compensador de atraso por el método de lugar geométrico de las raíces Ejemplo 2. Diseño de compensador de atraso por el método de lugar geométrico de las raíces Ejemplo 3. Diseño de compensador de atraso por el método de lugar geométrico de las raíces Ejemplo 4. Diseño de compensador de atraso por el método de

lugar geométrico de las raíces Ejemplo 5. Diseño de compensador de atraso por el método de lugar geométrico de las raíces 3. Diseño de compensador de adelanto - atraso por el método de lugar geométrico de las raíces Ejemplo 1. Diseño de compensador de adelanto - atraso por el método de lugar geométrico de las raíces Ejemplo 2. Diseño de compensador de adelanto - atraso por el método de lugar geométrico de las raíces Ejemplo 3. Diseño de compensador de adelanto - atraso por el método de lugar geométrico de las raíces

Electrónica de potencia Valor medio y Valor eficaz Detector de cruce por cero con optoacoplador El SCR 1. El optoscr 2. Rectificador monofásico de media onda controlado 3. Rectificador monofásico de onda completa semicontrolado 4. Rectificador monofásico de onda completa controlado 5. Rectificador trifásico de media onda controlado 6. Rectificador trifásico de onda completa semicontrolado 7. Rectificador trifásico de onda completa controlado El TRIAC

1. El optotriac 3. Controlador ac monofásico de onda completa El MOSFET 1. Troceador clase A

Adaptación de impedancias Máxima transferencia de potencia en dc Máxima transferencia de potencia en ac Transformación serie-paralelo y paralelo-serie de impedancias Redes de adaptación 1. Red de adaptación L 2. Red de adaptación Π 3. Red de adaptación T 4. Red de adaptación de dos o mas secciones L

Rectificador trifásico de onda completa semicontrolado

https://wilaebaelectronica.blogspot.com/2017/05/rectificador-trifasicode-onda-completa-semicontrolado.html

hola v1 1 0 sin(0 55 60 0 0 0) v2 2 0 sin(0 55 60 0 0 -120) v3 3 0 sin(0 55 60 0 0 -240) d1 8 1 d1n4004

d2 8 2 d1n4004 d3 8 3 d1n4004 .lib diode.lib R1 7 8 1k X1 1 4 7 2n1599 X2 2 5 7 2n1599 X3 3 6 7 2n1599 .lib THYRISTR.LIB Vg1 4 7 pulse(0 5 2.08m 1n 1n 10u 16.66m) Vg2 5 7 pulse(0 5 7.62m 1n 1n 10u 16.66m) Vg3 6 9 pulse(0 5 13.17m 1n 1n 10u 16.66m) Vg4 9 7 pulse(0 5 0.01m 1n 1n 10u 16.66m) .probe .tran 41u 16.66m 0 41u .end

ADICIONANDO UN DISPARADOR EN CERO PARA V3 hola v1 1 0 sin(0 55 60 0 0 0) v2 2 0 sin(0 55 60 0 0 -120) v3 3 0 sin(0 55 60 0 0 -240) d1 8 1 d1n4004 d2 8 2 d1n4004 d3 8 3 d1n4004 .lib diode.lib R1 7 8 1k

X1 1 4 7 2n1599 X2 2 5 7 2n1599 X3 3 6 7 2n1599 .lib THYRISTR.LIB Vg1 4 7 pulse(0 5 2.08m 1n 1n 10u 16.66m) Vg2 5 7 pulse(0 5 7.62m 1n 1n 10u 16.66m) Vg3 6 9 pulse(0 5 13.17m 1n 1n 10u 16.66m) Vg4 9 7 pulse(0 5 0.01m 1n 1n 10u 16.66m) ; son dos generadores de pulso en serie .probe .tran 41u 16.66m 0 41u .end

RECTIFICADOR TRIFASICO COMPLETAMENTE CONTROLADO ONDA COMPLETA CARGA R

hola v1 1 0 sin(0 55 60 0 0 0) v2 2 0 sin(0 55 60 0 0 -120) v3 3 0 sin(0 55 60 0 0 -240) x1 1 4 10 2n1599 x2 2 5 10 2n1599 x3 3 6 10 2n1599 r1 10 11 1k X4 11 7 1 2n1599

;X5 11 8 2 2n1599 X6 11 9 3 2n1599 .lib THYRISTR.LIB Vg1 4 10 pulse(0 5 3.46m 1n 1n 20u 16.66m) Vg2 5 10 pulse(0 5 9.01m 1n 1n 20u 16.66m) Vg3 6 10 pulse(0 5 14.56m 1n 1n 20u 16.66m) Vg4 7 1 pulse(0 5 11.8m 1n 1n 20u 16.66m) ;Vg5 8 2 pulse(0 5 0.69m 1n 1n 20u 16.66m) Vg6 9 3 pulse(0 5 6.25m 1n 1n 20u 16.66m) .probe D5 11 2 d1n4004 .lib diode.lib .tran 41u 16.66m 0 41u .end

IDA DISPARO DE X1 EN 60° X2, 120° DESPUÉS X3, 240° DESPUÉS

RETORNO DISPARO DE X5 EN 15°° DE X6, 120°DESPUÉS DE X4, 240° DESPUÉS

UNIDAD 4 DRIVERS, CHOPPERS-TROCEADORES DC/DC Y PROTECCIONES CONTROL PWM DE MOTOR DC

OPTO-ACOPLADOR SALIDA OPTTOSCR

DISPARO NO AISLADO

PROTECCIÓN CROWBAR EN LÍNEA DC

CONTROL RAMPA NIVEL DISPARO AISLDO

GENERACIÓN DE RAMPA SINCRÓNICA Y DISPARO AISLADOCOMPLETAR EL CIRCUITO-

CONVERSOR R-2R

CHOPPERS-TROCEADORES • Convertidor reductor (Buck): La tensión de salida es menor o igual que la tensión de entrada. E

• Convertidor elevador (Boost) : La tensión de salida en CD igual o superior a la tensión de entrada

• Convertidor reductor - elevador (Buck - Boost): La tensión de salida menor, igual o mayor que la tensión de entrada, pero con polaridad invertida

Tl494 Generador PWM De frecuencia hasta por los lados de los 200Kz y ancho de pulso variable. Recomendable >10kz Vcc 7a 40v Vin V(4) entre 0 y 2.50v para controlar entre 0% y 100% de ancho de pulso Los transistores de salida necesitan Rpullup Tambien genera una diente de sierra

Canal p

DRIVERS PARA INTERRUPTORES DE POTENCIA -CONT

PARA RELÉS

PARRA ICs

The TI TLC5940 is a 16 channel driver IC with 12 bit duty cycle PWM control (0 – 4095), 6 bit current limit control (0 – 63),

RFP30N06LE MOSFET transistor

the RFP30N06LE MOSFET transistor rated at 60V and 30A. This special MOSFET has a very low gate input voltage that works with 3.3V logic signals like those on mbed .

A typical MOSFET runs just a bit more efficiently if the gate input is a bit higher than the supply voltage. Special MOSFET driver ICs such as the LTC1155 use a charge pump circuit to drive the gate voltage higher on higher voltage MOSFET driver circuits

Any

The small breakout board seen above uses Toshiba’s TB6612FNG dual H-bridge motor driver IC, which can independently control two bidirectional DC motors or one bipolar stepper motor.

Stepper motor driver ICs Stepper motors have multiple coil windings that need to be activated in the current sequence to rotate the motor shaft by a fixed angle without the need for position feedback hardware.

Brushless DC motors Brushless DC motors are smaller and weigh less than equivalent DC (Brushed) motors, but they require a more complex speed control circuit. Brushless DC motors

▪

Optical Isolation Optical or mechanical isolation (relays) between higher voltage supplies and computers is always a good idea. Not having to

connect grounds between supplies helps in noisy industrial environments. Long wires can also pick up large transient voltages. seen below, the ILD2213T optocoupler IC used on the Sparkfun board contains two optically coupled pairs consisting of a Gallium Arsenide infrared LED and a silicon NPN phototransistor.

Power semiconductor devices are the heart of modern power electronics systems.

Power level requirements and switching frequency are increasing in the power electronics industry..

The gate of a MOSFET or IGBT is the electrically isolated control terminal for each device. drivers must be used to apply a voltage and provide drive current to the gate of the power device.

The gate driver circuit is an integral part of power electronics systems.

Gate drivers form an important interface between the high-power electronics and the control circuit and are used to drive power semiconductor devices.

Types of Gate Drivers Low-Side Drivers — Used to drive ground referenced switches (low side switches).

High-Side-Low-Side Drivers — Used to drive two switches connected in the bridge arrangement (both floating & ground referenced switches).

Gate Driver Isolation Gate drive circuits for power inverters and converters often require electrical isolation

In a simple bridge topology structure with a driving circuit as shown here, the source terminal of switch 1 can be floating anywhere from ground to DC bus potential. Therefore two things are needed for driving high side switches: 1. Floating supply — to provide power to any circuitry associated with this floating midpoint potential.

2. Level shifter — to convey the PWM control signal to the floating driver circuitry.

Isolation Techniques gate drive transformers.

Author

•

•

Bhuvana Madhaiyan Bhuvana Madhaiyan is a Design and Development Engineer with Talema India. She holds a bachelor’s degree in Electrical and Electronics Engineering from Anna University

TRABAJO GRUPAL 1, CARGA R, DISPARO AISLADO DE SCR,, ENTRADA DE CONSIMNAS, TIEMPO IDA, TIEMPO REGRESO POR TECLADO DE PC. SEPTIEMBRE 22 INICIO

CHEQUEO ENCENDIDO-APAGADO

An Introduction to Thyristors and Their Applications M. Ramamoorty Professor of Electrical Engineering Indian Institute of Technology Kanpur file:///C:/Users/PEDRO/Downloads/1977_Bookmatter_AnIntroductionT oThyristorsAndT.pdf

Published in the United Kingdom 1978 by THE MACMILLAN PRESS LIMITED London and Basingstoke Associated comppnies in Delhi Dublin Hong Kong Johannesburg Lagos Melbourne New York Singapore and Tokyo

DRIVERS AND PROTECTIONS-CONTINUATIONS

High Frequency Floating Power Supply

UNIDADES 5 ,6 CONVERSIÓN DC/AC, CIRCUITOS DE PROTECCIÓN PROTECCIONES POR OPERACIÓN ADECUADA SEGURA

PROTECCIONES PARA EXCITACIÓN DC

Principal Auxiliar 1 Auxiliar 2 Auxiliar 3 C4 C3 C2 C1 C4 C3 C2 C1 C4 C3 C2 C1 C4 C3 C2 C1 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 0 1 0 1 1 0 0 1 1 0 0 1 1 0 1 0 0 1 0 1 1 0 0 1 1 0 0 1 0 1 0 1 1 0 0 1 1 0 0 1 1 0 0 1 0 1 0 1 0 1 0 1 1 0 0 1 1 0 0 1 0 1 1 0 0 1 0 1 1 0 0 1 1 0 1 0 0 1 0 1 1 0 0 1 1 0 0 1 1 0 1 0 0 1 0 1 0 1 0 1 1 0 0 1 1 0 1 0 0 1 1 0 0 1 0 1 1 0 0 1 0 1 0 1 1 0 0 1 1 0 0 1 1 0 0 1 0 1 0 1 1 0 0 1 0 1 0 1 1 0 0 1 0 1 0 1 1 0 1 0 0 1 0 1 1 0 0 1 0 1 0 1 0 1 0 1 1 0 0 1 1 0 0 1 0 1 0 1 0 1 0 1 0 1 0 1 1 0 0 1 0 1 0 1 0 1 1 0 0 1 0 1 1 0 0 1 0 1 1 0 0 1 0 1 1 0 0 1 1 0 0 1 0 1 1 0 0 1 0 1 0 1 0 1 1 0 0 1 0 1 1 0 0 1 1 0 0 1 0 1 1 0 1 0 0 1 0 1 1 0 0 1 1 0 0 1 1 0 1 0 0 1 0 1 1 0 0 1 0 1 0 1 1 0 1 0 0 1 0 1 1 0 1 0 0 1 0 1 1 0 1 0 0 1 0 1 0 1 0 1 1 0 0 1 1 0 1 0 0 1 0 1 0 1 0 1 0 1 0 1 1 0 1 0 0 1 0 1 0 1 1 0 0 1 0 1 1 0 1 0 0 1 1 0 0 1 0 1 1 0

Voltaje -93,33 -91 -88,67 -86,33 -84 -81,67 -79,33 -77 -74,67 -72,33 -70 -67,67 -65,33 -63 -60,67 -58,33 -56 -53,67 -51,33 -49 -46,67 -44,33 -42 -39,67 -37,33

0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0

1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1

1 1 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0

0 0 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1

1 1 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1

0 0 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 0 0 0 0 0 0 0

0 0 1 1 1 1 1 1 1 1 1 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0

1 1 0 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1

1 1 0 0 0 0 0 0 1 1 1 0 0 0 0 0 0 1 1 1 0 0 0 0 0 0 1

0 0 1 1 1 1 1 1 0 0 0 1 1 1 1 1 1 0 0 0 1 1 1 1 1 1 0

0 0 1 1 1 0 0 0 0 0 0 1 1 1 0 0 0 0 0 0 1 1 1 0 0 0 0

1 1 0 0 0 1 1 1 1 1 1 0 0 0 1 1 1 1 1 1 0 0 0 1 1 1 1

0 1 0 0 1 0 0 1 0 0 1 0 0 1 0 0 1 0 0 1 0 0 1 0 0 1 0

1 0 1 1 0 1 1 0 1 1 0 1 1 0 1 1 0 1 1 0 1 1 0 1 1 0 1

0 0 1 0 0 1 0 0 1 0 0 1 0 0 1 0 0 1 0 0 1 0 0 1 0 0 1

1 1 0 1 1 0 1 1 0 1 1 0 1 1 0 1 1 0 1 1 0 1 1 0 1 1 0

-35 -32,67 -30,33 -28 -25,67 -23,33 -21 -18,67 -16,33 -14 -11,67 -9,33 -7 -4,67 -2,33 0 2,33 4,67 7 9,33 11,67 14 16,33 18,67 21 23,33 25,67

0 0 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1

1 1 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0

0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0

1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1

1 1 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1

0 0 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 0 0 0 0 0 0 0

0 0 1 1 1 1 1 1 1 1 1 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0

1 1 0 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1

1 1 0 0 0 0 0 0 1 1 1 0 0 0 0 0 0 1 1 1 0 0 0 0 0 0 1

0 0 1 1 1 1 1 1 0 0 0 1 1 1 1 1 1 0 0 0 1 1 1 1 1 1 0

0 0 1 1 1 0 0 0 0 0 0 1 1 1 0 0 0 0 0 0 1 1 1 0 0 0 0

1 1 0 0 0 1 1 1 1 1 1 0 0 0 1 1 1 1 1 1 0 0 0 1 1 1 1

0 1 0 0 1 0 0 1 0 0 1 0 0 1 0 0 1 0 0 1 0 0 1 0 0 1 0

1 0 1 1 0 1 1 0 1 1 0 1 1 0 1 1 0 1 1 0 1 1 0 1 1 0 1

0 0 1 0 0 1 0 0 1 0 0 1 0 0 1 0 0 1 0 0 1 0 0 1 0 0 1

1 1 0 1 1 0 1 1 0 1 1 0 1 1 0 1 1 0 1 1 0 1 1 0 1 1 0

28 30,33 32,67 35 37,33 39,67 42 44,33 46,67 49 51,33 53,67 56 58,33 60,67 63 65,33 67,67 70 72,33 74,67 77 79,33 81,67 84 86,33 88,67

1 1

0 0

0 0

1 1

1 1

0 0

0 0

1 1

1 1

0 0

0 0

1 1

0 1

1 0

0 0

1 1

91 93,33

https://www.youtube.com/watch?v=AXhupj2gE5I Driver.

El driver es un elemento encargado de adaptar la señal de control de los MOSFET a la señal adecuada para que el transistor trabaje en zona óhmica o corte.

Se puede afirmar que

un driver debe cumplir tres objetivos: Proporcionar una entrada de alta impedancia respecto al circuito de control.

Adaptar niveles de tensión.

Driver.

El driver es un elemento encargado de adaptar la señal de control de los MOSFET a la señal adecuada para que el transistor trabaje en zona óhmica o corte. Se puede afirmar que un driver debe cumplir tres objetivos: Proporcionar una entrada de alta impedancia respecto al circuito de control.

Driver para el disparo de un MOSFET con su efecto parásito más Capacidad de corriente de salida ig elevada. Esto permite realizar la conmutación rápidamente, pues influye en la velocidad de carga del condensador parásito equivalente Ciss.

Adicionalmente, los drivers comerciales pueden incorporar tiempos muertos y protecciones apropiados para algunas aplicaciones.

Existen diferentes opciones a la hora de seleccionar un driver. Se pueden emplear componentes discretos [21] para diseñar un driver,

optoacopladores o drivers fabricados como circuitos integrados. Cada uno de ellos tiene unas ventajas e inconvenientes.

Para el diseño se ha utilizado un circuito integrado IR2110 del fabricante International Rectifier. Es un Driver para MOSFET de canal N de alta velocidad e IGBTs. Se compone como se muestra en la figura 4.13 de dos drivers independientes, el del lado bajo (Mosfet referido a masa) y el del lado alto (Mosfet no referido a masa). Tiene un buffer de entrada de alta impedancia y entrada para deshabilitar el driver. También incorpora una adaptación de niveles para el disparo del mosfet no referido a masa a partir de una tensión de bootstrap.

Por último, una etapa de salida de alta corriente permite el encendido/apagado rápido de los interruptores minimizando las pérdidas en conmutación. Soporta tensiones en la etapa de salida de hasta 500V .importante en el disparo CGS capacidad puerta-fuente. A

daptar niveles de tensión.

DRIVER. 37 En la figura 4.14 se presenta un conexionado típico del circuito adquirido de la hoja de características del componente. A la vista de la figura, habrá que emplear un integrado IR2110 para cada rama del puente H. VDD es la tensión de alimentación de la parte lógica del circuito. La parte de entrada del circuito.

HIN es la entrada que comanda la parte alta del driver y cuya salida en fase es HO. LIN es la entrada que comanda la parte baja del driver y cuya salida en fase es LO.

SD es una entrada de control de apagado del driver, similar a un enable.

V ss es la masa del circuito lógico.

VB es una entrada de tensión flotante referida a Vs para el disparo del MOSFET de la parte alta.

Vcc es la entrada de tensión referida a COM para el disparo del M

OSFET de la parte baja. Vcc no es flotante.

Alta impedancia de entrada.

El driver IR2110 presenta una entrada de alta impedancia (ver figura 4.15) con una corriente de polarización máxima de 40_A. La entrada dispone de un comparador con histéresis de modo que lo hace muy

robusto al ruido que pueda estar presente en las señales de control.

Adaptación de niveles.

El circuito de control genera los disparos con tensiones lógicas entre 0V y 5V , sin embargo, dos factores impiden que estas tensiones sean aplicadas directamente a la puerta del MOSFET. En primer lugar, a la vista de la figura 4.7, se observa que una tensión puerta-fuente de 5V es insuficiente para lograr la saturación del MOSFET lo que daría lugar a elevadas pérdidas en conducción.

En segundo lugar, los MOSFET de la parte alta (Los interruptores que durante todo el libro se han llamado S1 y S2) no tienen su puerta unida a masa, por lo que es necesaria una manera de aplicar una tensión flotante para dispararlos. Existen diferentes soluciones que se presentarán a continuación Adaptación de niveles referidos a masa.

Los MOSFET de la parte baja, S3 y S4, al estar referidos a masa solo necesita de la adaptación de niveles de tensión. El Driver IR2110 tiene una doble alimentación, una para la lógica VDD 5V y otra para la alimentación del circuito de disparo Vcc 15V . La figura 4.16 el diagrama de bloques simplificado del integrado con una salida en MOSFET complementario.

Adaptación de niveles no referidos a masa.

Los MOSFET de la parte alta S1 y S2 no están referidos a masa. Por ello no es suficiente realizar un cambio de nivel de tensión sino que además esta tensión debe tener su masa flotante. Es necesario conectar una fuente de tensión flotante de 15V en VB referida a VS. Existen diferentes alternativas para generar esta tensión:

Condensador de Bootstrap. Utilizar un condensador de Bootstrap es la opción que se presenta en la nota de aplicación de International Rectifier AN-978 [10] y en el consejo de aplicación DT 98-2a[3].

Consiste en conectar un condensador Cbs y un diodo Dbs como se muestra en la figura 4.18, de manera que cuando el MOSFET de la parte baja conduce, el condensador se carga a través del diodo con la tensión Vcc. Cuando el MOSFET de la parte baja deja de conducir, el diodo evita que el condensador se descargue. Si la capacidad del condensador es suficiente, este proporcionará una tensión flotante estable para el encendido del MOSFET superior.

En [3] se presenta una fórmula para el cálculo de la capacidad de Cbs en función de la capacidad de puerta del MOSFET, la frecuencia de conmutación y otros parámetros como las corrientes de pérdidas existentes, caídas de tensión, etc.

Convertidor aislado. Una opción sencilla y eficaz es emplear un convertidor comercial con aislamiento para generar esta tensión como se muestra en la figura 4.19. La tensión de aislamiento necesaria es de al menos Vdc, que es la tensión continua de potencia y se ha decidido serán 320V . La potencia mínima del convertidor se puede cal cular con las siguientes fórmulas: Como el periodo de conmutación es mucho mayor que la constante de tiempo del sistema formado por RG y Cgs (condensador parásito de puerta del mosfet). La energía necesariamplear un condensador de otstrap o emplear un pequeño convertidor aislado que genere esta tensión

eatures Product Summary n Floating channel designed for bootstrap operation Fully operational to +600V Tolerant to negative transient voltage dV/dt immune n Gate drive supply range from 10 to 20V n Undervoltage lockout for both channels n Separate logic supply range from 5 to 20V Logic and power ground ±5V offset n CMOS Schmitt-triggered inputs with pull-down n Cycle by cycle edge-triggered shutdown logic n Matched propagation delay for both channels n Outputs in phase with inputs Description V OFFSET I +/O V OUT t on/off (typ.)

Delay Matching Packages 600V max.

2A / 2A 10 - 20V 120 & 94 ns 10 ns The IR2113 is a high voltage, high speed power MOSFET and IGBT driver with independent high and low side referenced output channels. Proprietary

HVIC and latch immune CMOS technologies enable ruggedized monolithic construction. Logic inputs are compatible with standard CMOS or LSTTL outputs.

The output drivers feature a high pulse current buffer stage designed for minimum driver cross-conduction. Propagation delays are matched to simplify use in high frequency applications. The floating channel can be used to drive an N-channel power MOSFET or IGBT in the high side configuration which operates up to 600 volts.

Typical Connection IR2113 CONTROLADOR DE PUERTA DE MEDIO PUENTE IC NO INVERTIR El IR2113 es un poder de alta tensión y alta velocidad Controlador MOSFET e IGBT con alto independiente y canales de salida con bajo nivel de referencia. Propiedad Las tecnologías HVIC y latch inmune CMOS permiten construcción monolítica reforzada. Las entradas lógicas son compatible con salidas estándar CMOS o LSTTL. Los controladores de salida con un alto buffer de corriente de pulso etapa diseñada para la conducción cruzada mínima del conductor.

Los retrasos de propagación se combinan para simplificar uso en aplicaciones de alta frecuencia.

El flotante el canal se puede usar para impulsar una potencia de canal N MOSFET o IGBT en la cámara de lado alto que

opera

hasta

600

CARACTERISTICAS: Canal flotante diseñado para la operación de arranque Totalmente operativo a + 500 V o + 600 V Tolerante a voltaje transitorio negativo dV / dt inmune Rango de suministro del controlador de compuerta de 10 a 20V Bloqueo de baja tensión para ambos canales 3.3V compatible con lógica Rango de suministro lógico separado de 3.3V a 20V Tierra lógica y de potencia ± 5V offset Entradas CMOS activadas por Schmitt con menú desplegable

voltios.

Ciclo por ciclo de lógica de apagado por disparo de borde Retardo de propagación coincidente para ambos canales Salidas en fase con entradas ESPECIFICACIONES TECNICAS: Paquete / Cubierta: PDIP-14 Producto: Half-Bridge Drivers Número de controladores: 2 Driver Corriente de salida: 2.5 A Configuración: Non-Inverting Tiempo de subida: 35 ns Tiempo de caída: 25 ns Voltaje de alimentación - Máx.: 20 V Voltaje de alimentación - Mín.: 10 V Retraso de propagación - Máx.: 150 ns Temperatura de trabajo mínima: - 40 C Temperatura de trabajo máxima: + 125 C

Referencia de producto: IR2113PBF DRIVER

UNIDAD 6 PROTECCIONES

(di/dt)@>>Vcc/L Protección contra crecimientos de la corriente

(dVak/dt)>>Vcc*R/L Protección contra velocidad de crecimiento del voltaje AK

R1=100 isminuye la sensibilidad de Ig C1=0.1uf Envía transitorios de Gate a tierra R2 Limita la IGT y la IZ

SIN DIODO RUEDALIBRE

Vim=156V

hola v1 1 0 sin(0 156 60 0 0 0) x1 1 2 3 2n1599 .LIB THYRISTR.lib r1 3 4 1k l1 4 0 50m ic=0 vg1 2 3 pulse(0 5 2.78m 1n 1n 10u 16.66m) .probe .tran 41u 16.66m 0 41u uic .end

CON DIODO RUEDA LIBRE

Vim=156V

hola v1 1 0 sin(0 156 60 0 0 0) x1 1 2 3 2n1599 .LIB THYRISTR.lib d1 0 3 d1n4004 .lib diode.lib

r1 3 4 1k l1 4 0 50m ic=0 vg1 2 3 pulse(0 5 2.78m 1n 1n 10u 16.66m) .probe .tran 41u 16.66m 0 41u uic .end

RECTIFICADOR MONOFÁSICO NO CONTROLADO CARGA R,L,E

RECTIFICADOR MONOFÁSICO CONTROLADO CON CARGA RLE

RECTIFICADOR SEMICONTROLADO MONOFÁSDICO CON CARGA RL

RECTIFICADOR SEMICONTROLADO MONOFÁSDICO CON CARGA RLE

Published in the United Kingdom 1978 by THE MACMILLAN PRESS LIMITED London and Basingstoke Associated comppnies in Delhi Dublin Hong Kong Johannesburg Lagos Melbourne New York Singapore and Tokyo

DRIVER BOOSTRAP

En el dominio de los circuitos MOSFET, el” Bootstrapping” se usa comúnmente para levantar el punto de operación de un transistor por encima del riel de la fuente de alimentación

https://bibdigital.epn.edu.ec/bitstream/15000/19390/1/CD-8776.pdf

https://bibdigital.epn.edu.ec/bitstream/15000/19390/1/CD-8776.pdf

Power generation, transmission, distribution, and electric machines in india In 1960 course in industrial electronics with thyristor converter The silicon-based power

devices, such as thyristors, triacs, GTOs, bipolar-junction transistors (BJTs), power MOSFETS, IGBTs, and IGCTs, have already established their applications. Thyristor- (and diode-) based phase-controlled converters and cycloconverters, which established the modern solid-state power electronics age, are still used extensively on utility systems and cause power quality and power factor problems.

Different harmonic standards have been introduced to combat the power quality (PQ) problems. The traditional phase control can be replaced by self-controlled devices (MOSFETS, IGBTs, and IGCTs) at PWM to solve the aforementioned problems.

Alternatively, harmonic filters (active, passive, or hybrid) can be used to

solve the PQ problems, and SVCs can solve the PF problem. Such a solution is sometimes justified as a retroactive solution to an expensive installation.

The phase-controlled devices will eventually fall into obsolescence in favor of the self-controlled devices.

The BJTs are already obsolete, where the lower end has been absorbed by power MOSFETs and the higher end by IGBTs. Insulated-gate MOSFETs and IGBTs have now established their importance covering from low-power to high-power applications.

Again, the GTOs tha 1956, the ‘‘PNPN triggering transistor’’ was invented and commercially introduced as a silicon-controlled rectifier (SCR) or

thyristor (solid-state thyratron) by General Electric (GE) in 1958. Thi

Silicon power diodes started appearing in 1956

Toshiba also introduced giant transistors. Power MOSFET was developed around the same time in 1978 SPICE, “Simulation program with integrated emphasis” fue desarrollado grupalmente en 1972 en Berkeley, Universidad de California Estados Unidos, por estudiantes de pregrado, de Doctorado en Electrónica, y post Doctorado en Matemáticas, Física, Química y expertos en programación de computadoresde código fuente.

Profesor Donald O. Pederson el 12 de Abril de 1973, el simulador SPICE en el “Sixteenth Midwest Symposium on Circuit Theory“

“Donald Oscar Pederson Profesor de Ingeniería Eléctrica e Informática, Emérito.

Berkeley 1925— 2004”

"Nunca esperes la aprobación, no le digas a nadie que estás haciendo algo, solo hazlo", dijo Pederson más tarde. "Ese es mi lema."

Tomado de https://senate.universityofcalifornia.edu/_files/inmemoriam/html/donald oscarpederson.htm La estructura de un programa en SPICE texto es:

Estructura de de prorama SPICE texto

La primera línea es comentario y si se necesitan otros, sus líneas se hacen preceder con ; o

Características principales:

Analiza diferentes aplicaciones y su control.

Explica los métodos de diseño de controladores más importantes, tanto en analógico como digital.

Describe aplicaciones diferentes, pero importantes, que se pueden utilizar en futuros productos industriales.

Cubre convertidores de fuente de voltaje con gran detalle Demuestra aplicaciones en un contexto mucho más amplio.

Tabla de contenido:

La electrónica de potencia es la aplicación de la electrónica de estado sólido para controlar y convertir una forma de energía eléctrica en otra,

como convertir entre CA y CC o cambiar la magnitud y fase del voltaje y la corriente o frecuencia o una combinación de estos. Se presentan varios convertidores electrónicos de potencia (CA-CC, CC-CA, CC-CC y CA-CA) y los circuitos y topologías de uso común con su principio de funcionamiento general.

VARIACIÓN DE PARÁMETROS EN SPICE DISEÑO ASISTIDO

FÓRMULA SEMILLA C=Io*Δt/Vriz Deseamos Io=0.3A DC Δt 8.33m C=Vriz=3V C=833uf aproximadamente Lo que hacemos serávariar este capacitor alrededor de este valor

Hola V1 1 2 sin(0 50 60 0 0 0) D1 1 3 d1n4004 D2 2 3 d1n4004 D3 0 1 d1n4004 D4 0 2 d1n4004 .Lib diode.lib R1 3 0 100 C1 3 0 {cvari} ic=41 .param cvari=24u .step param cvari 20u 470u 20u

.probe .tran 41u 16.66m 0 41u uic .end

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The Basics of Power Semiconductor Devices: Structures, Symbols, and Operations

TECHNICAL ARTICLE The Basics of Power Semiconductor Devices: Structures, Symbols, and Operations June 13, 2015 by Editorial Team This technical article is dedicated to the review of the following power electronics devices which act as solid-state switches in the circuits.

These act as a switch without any mechanical movement.

The Structures, Electronic Symbols, Basic Operations and Several Characteristics Representations of Power Semiconductor Devices Recommended Level Beginner Power Electronic Devices That Act As Solid-State Switches This technical article is dedicated to the review of the following power electronic devices which act as solid-state switches in the circuits. They act as a switch without any mechanical movement.

• • •

Power Diodes Metal-Oxide-Semiconductor Field-Effect (MOSFET) Bipolar -Junction Transistor (BJT)

Transistor

• •

Insulated-Gate Bipolar Transistor (IGBT) Thyristors (SCR, GTO, MCT)

Solid-state devices are completely made from a solid material and their flow of charges is confined within this solid material. This name “solid state” is often used to show a difference with the earlier technologies of vacuum and gas-discharge tube devices; and also to exclude the conventional electro-mechanical devices (relays, switches, hard drives and other devices with moving parts).

The transistor by Bell Labs in 1947 was the first solid-state device to come into commercial use later in the 1960s. In this article, similar solidstate devices such as power diode, power transistor, MOSFET, thyristor and its two-transistor model, triac, gate turn-off thyristor (GTO), insulated-gate bipolar transistor (IGBT) and their characteristics (such as i-v characteristics and turn-off characteristics) is also presented. In power electronics circuitry, these switches act in saturation region and work in linear region in the analog circuitry such as in power amplifiers and linear regulators. This makes these switches highly efficient since there are lesser losses during the power processing.

Power Diodes A power diode has a P-I-N structure as compared to the signal diode having a P-N structure. Here, I (in P-I-N) stands for intrinsic semiconductor layer to bear the high-level reverse voltage as compared to the signal diode (n- , drift region layer shown in Fig. 2). However, the drawback of this intrinsic layer is that it adds noticeable resistance during forward-biased condition. Thus, power diode requires a proper cooling arrangement for handling large power dissipation. Power diodes are

used in numerous applications including rectifier, voltage clamper, voltage multiplier and etc. Power diode symbol is the same as of the signal diode as shown in Fig.1.

Figure 1. Symbol for Power Diode

Figure 2. Structure of Power Diode

Other features that are incorporated in the power diode letting it handle larger power are: (a) Use of guard rings (b) Coating of Silicon Dioxide layer

Guard rings are of p-type that prevents their depletion layer merge with the depletion layer of the reverse-biased p-n junction. The guard rings prevent the radius of the curvature of the depletion layer boundary to become too narrow which increases the breakdown strength. Coating of SiO2 layer helps in limiting the electric field at the surface of the power diode.

If thickness of lightly doped I layer (n- layer) > depletion layer width at breakdown ⇒ Non-punch through power diode.

(This means depletion layer has not punched through the lightly-doped n-layer.)

If thickness of I layer < depletion layer width at breakdown ⇒ Punch through power diode.

Characteristics of Power Diode The two types of characteristics of a power diode are shown in Fig. 3 and Fig. 4 named as follows: (i) Amp-volt characteristics (i-v characteristics) (ii) Turn-off characteristics (or reverse-recovery characteristics)

Figure 3. Amp-Volt Characteristics of Power Diode

Cut-in voltage is the value of the minimum voltage for VA (anode voltage) to make the diode works in forward conducting mode. Cut-in voltage of signal diode is 0.7 V while in power diode it is 1 V. So, its typical forward conduction drop is larger. Under forward-bias condition, signal diode current increases exponentially and then increases linearly. In the case of the power diode, it almost increases linearly with the applied voltage as all the layers of P-I-N remain saturated with minority carriers under forward bias. Thus, a high value of current produces results in voltage drop which mask the exponential part of the curve. In reverse-bias

condition, small leakage current flows due to minority carriers until the avalanche breakdown appears as shown in Fig. 3.

Figure 4. Turn-Off Characteristics of Power Diode: a) Variation of Forward Current if ; b) Variation of Forward Voltage Drop vf ; c) Variation of Power Loss

After the forward diode comes to null, the diode continues to conduct in the opposite direction because of the presence of stored charges in the depletion layer and the p or n-layer. The diode current flows for a reverse-recovery time trr. It is the time between the instant forward diode current becomes zero and the instant reverse-recovery current decays to 25 % of its reverse maximum value.

Time Ta : Charges stored in the depletion layer removed.

Time Tb : Charges from the semiconductor layer is removed.

Shaded area in Fig 4.a represents stored charges QR which must be removed during reverse-recovery time trr.

Power loss across diode = vf * if

(shown in Fig. 4.c)

As shown, major power loss in the diode occurs during the period t b.

Recovery can be abrupt or smooth as shown in Fig. 5. To know it quantitatively, we can use the S – factor.

Ratio Tb/Ta : Softness factor or S-factor.

S-factor: measure of the voltage transient that occurs during the time the diode recovers.

S-factor = 1 ⇒ low oscillatory reverse-recovery process. (Soft –recovery diode) S-factor <1 ⇒ large oscillatory over voltage (snappy-recovery diode or fast-recovery diode).

Power diodes now exist with forward current rating of 1A to several thousand amperes with reverse-recovery voltage ratings of 50V to 5000V or more.

Figure 5. Reverse-Recovery Characteristics for Power Diode

Schottky Diode: It has an aluminum-silicon junction where the silicon is an n-type. As the metal has no holes, there is no stored charge and no reverse-recovery time. Therefore, there is only the movement of the majority carriers (electrons) and the turn-off delay caused by recombination process is avoided. It can also switch off much faster than a p-n junction diode. As compared to the p-n junction diode it has: (a) Lower cut-in voltage (b) Higher reverse leakage current

(c) Higher operating frequency Application: high-frequency instrumentation and switching power supplies.

Figure 6. Schottky Diode Symbol and Current-Voltage Characteristics Curve

Metal-Oxide Semiconductor Field-Effect Transistor (MOSFET) MOSFET is a voltage-controlled majority carrier (or unipolar) threeterminal device. Its symbols are shown in Fig. 7 and Fig. 8. As compared to the simple lateral channel MOSFET for low-power signals, power MOSFET has different structure. It has a vertical channel structure

where the source and the drain are on the opposite side of the silicon wafer as shown in Fig. 10. This opposite placement of the source and the drain increases the capability of the power MOSFET to handle larger power.

Figure 7. MOSFET Symbol

Figure 8. MOSFET Symbols for Different Modes

In all of these connections, substrates are internally connected. But in cases where it is connected externally, the symbol will change as shown in the n-channel enhancement type MOSFET in Fig. 9. N-channel enhancement type MOSFET is more common due to high mobility of electrons.

Figure 9. N-channel Enhancement-Type MOSFET with Substrate Connected Externally

Figure 10. Cross-Sectional View of the Power MOSFET

Basic circuit diagram and output characteristics of an n-channel enhancement power MOSFET with load connected are in Fig. 11 and Fig. 12 respectively.

Figure 11. Power MOSFET Structural View with Connections

Drift region shown in Fig. 11 determines the voltage-blocking capability of the MOSFET. When VGS = 0, ⇒ VDD makes it reverse biased and no current flows from drain to source.

When VGS > 0, ⇒ Electrons form the current path as shown in Fig. 11. Thus, current from the drain to the source flows. Now, if we will increase the gate-tosource voltage, drain current will also increase.

Figure 12. Drain Current (ID) vs Drain-to-Source Voltage (VDS) Characteristics Curves

For lower value of VDS, MOSFET works in a linear region where it has a constant resistance equal to VDS / ID. For a fixed value of VGS and greater than threshold voltage VTH, MOSFET enters a saturation region where the value of the drain current has a fixed value.

Figure 13. Output Characteristics with Load Line

If XY represents the load line, then the X-point represents the turn-off point and Y-point is the turn-on point where VDS = 0 (as voltage across the closed switch is zero). The direction of turning on and turning off process is also shown in Fig. 13.

Besides the output characteristics curves, transfer characteristics of power MOSFET is also shown in Fig. 14.

Figure 14. Gate-to-Source Voltage vs. Drain Current Characteristics for Power MOSFET

Here, VTH is the minimum positive voltage between gate and the source above which MOSFET comes in on-state from the off-state. This is called threshold voltage. It is also shown in the output characteristics curve in Fig. 12.

Close view of the structural diagram given in Fig. 11 reveals that there exists a fictitious BJT and a fictitious diode structure embedded in the power MOSFET as shown in Fig. 15.

As source is connected to both base and emitter of this parasitic BJT, emitter and base of the BJT are short circuited. That means this BJT acts in cut-off state.

Figure 15. Fictitious BJT and Fictitious Diode in the Power MOSFET

Fictitious diode anode is connected to the source and its cathode is connected to the drain. So, if we apply the negative voltage V DD across the drain and source, it will be forward biased. That means, the reverseblocking capability of the MOSFET breaks. Thus, this can be used in inverter circuit for reactive loads without the need of excessive diode across a switch. Symbolically, it is represented in Fig. 16.

Figure 16. MOSFET Representation with Internal Body Diode

Although MOSFET internal body diode has sufficient current and switching speed for most of the applications, there may be some applications where the use of ultra-fast diodes is required. In such cases, an external fast-recovery diode is connected in an antiparallel manner.

But a slow-recovery diode is also required to block the body diode action as given in Fig. 17.

Figure 17. Implementation of Fast-Recovery Diode for Power MOSFET

One of the important parameters that affects the switching characteristics is the body capacitances existing between its three terminals i.e. drain, source and gate. Its representation is shown in Fig. 18.

Figure 18. MOSFET Representation Showing Junction Capacitances

Parameters CGS, CGD and CDS are all non-linear in nature and given in the device’s data sheet of a particular MOSFET. They also depend on the DC bias voltage and the device’s structure or geometry. They must be charged through gate during turn-on process to actually turn on the

MOSFET. The drive must be capable of charging and discharging these capacitances to switch on or switch off the MOSFET. Thus, the switching characteristics of a power MOSFET depend on these internal capacitances and the internal impedance of the gate drive circuits. Also, it depends on the delay due to the carrier transport through the drift region. Switching characteristics of power MOSFET are shown in Fig. 19 and Fig. 20.

Figure 19. Turn-on Characteristics of Power MOSFET

There is a delay from t0 to t1 due to charging of input capacitance up to its threshold voltage VTH. Drain current in this duration remains at zero value. This is called a delay time. There is a further delay from t 1 to t2 during which the gate voltage rises to V GS, a voltage required to drive the MOSFET into on-state. This is called the rise time. This total delay can be reduced by using a low-impedance drive circuit. The gate current during this duration decreases exponentially as shown. For the time greater than t2, the drain current ID has reached its maximum constant value I. As drain current has reached the constant value, the gate-tosource voltage is also constant as shown in the transfer characteristics of MOSFET in Fig. 20.

Figure 20. Transfer Characteristics of Power MOSFET with Operating Point

For turn-off characteristics, assume that the MOSFET is already in the switched-on situation with steady state. As t = t 0, gate voltage is reduced to zero value; CGS and CGD start to discharge through gate resistance RG. This causes a turn-off delay time up to t1 from t0 as shown in Fig. 21.

Assuming the drain-to-source voltage remains fixed. During this duration, both VGS and IG decreases in magnitude, drain current remains at a fixed value drawing current from CGD and CGS.

Figure 21. Turn-Off Characteristics of Power MOSFET

For the time where t2 > t > t1, gate-to-source voltage is constant. Thus, the entire current is now being drawn from CGD. Up to time t3, the drain current will almost reach zero value; which turns off the MOSFET. This time is known as the fall time, this is when the input capacitance discharges up to the threshold value. Beyond t 3, gate voltage decreases exponentially to zero until the gate current becomes zero.

Power Bipolar Junction Transistor (BJT) Power BJT is used traditionally for many applications. However, IGBT (Insulated-Gate Bipolar Transistor) and MOSFET (Metal-OxideSemiconductor Field-Effect Transistor) have replaced it for most of the applications but still they are used in some areas due to its lower saturation voltage over the operating temperature range. IGBT and MOSFET have higher input capacitance as compared to BJT. Thus, in case of IGBT and MOSFET, drive circuit must be capable to charge and discharge the internal capacitances.

Figure 22. (a) NPN BJT (b) PNP BJT

The BJT is a three-layer and two-junction npn or pnp semiconductor device as given in Fig. 22. (a) and (b).

Although BJTs have lower input capacitance as compared to MOSFET or IGBT, BJTs are considerably slower in response due to low input impedance. BJTs use more silicon for the same drive performance.

In the case of MOSFET studied earlier, power BJT is different in configuration as compared to simple planar BJT. In planar BJT, collector and emitter is on the same side of the wafer while in power BJT it is on the opposite edges as shown in Fig. 23. This is done to increase the power-handling capability of BJT.

Figure 23. Power BJT PNP Structure

Power n-p-n transistors are widely used in high-voltage and high-current applications which will be discussed later.

Input and output characteristics of planar BJT for common-emitter configuration are shown in Fig. 24. These are current-voltage characteristics curves.

Figure 24. Input Characteristics and Output Characteristics for the Common-Emitter Configuration of Planar BJT respectively

Characteristic curves for power BJT is just the same except for the little difference in its saturation region. It has additional region of operation known as quasi-saturation as shown in Fig. 25.

Figure 25. Power BJT Output Characteristics Curve

This region appears due to the insertion of lightly-doped collector drift region where the collector base junction has a low reverse bias. The resistivity of this drift region is dependent on the value of the base current. In the quasi-saturation region, the value of ß decreases significantly. This is due to the increased value of the collector current with increased temperature. But the base current still has the control over the collector current due to the resistance offered by the drift region. If the transistor enters in hard saturation region, base current has no control over the collector current due to the absence of the drift region and mainly depends on the load and the value of VCC. A forward-biased p-n junction has two capacitances named depletion layer capacitance and diffused capacitance. While a reverse bias junction has only a depletion capacitance in action. Value of these capacitances depends on the junction voltage and construction of the transistor. These capacitances come into role during the transient

operation i.e. switching operations. Due to these capacitances, transistor does not turn on or turn off instantly.

Switching characteristics of power BJT is shown in Fig.26. As the positive base voltage is applied, base current starts to flow but there is no collector current for some time. This time is known as the delay time (td) required to charge the junction capacitance of the base to emitter to 0.7 V approx. (known as forward-bias voltage). For t > t d, collector current starts rising and V CE starts to drop with the magnitude of 9/10th of its peak value. This time is called rise time, required to turn on the transistor. The transistor remains on so long as the collector current is at least of this value.

For turning off the BJT, polarity of the base voltage is reversed and thus the base current polarity will also be changed as shown in Fig. 26. The base current required during the steady-state operation is more than that required to saturate the transistor. Thus, excess minority carrier charges are stored in the base region which needs to be removed during the turnoff process. The time required to nullify this charge is the storage time, ts. Collector current remains at the same value for this time. After this, collector current starts decreasing and base-to-emitter junction charges to the negative polarity; base current also get reduced.

ypes of circuit protection 1.

2.

3.

4.

5.

Fuse Residual Current Device (RCD) Miniature Circuit Breaker (MCB) Residual Current Breaker with Overcurrent (RCBO) Protection Relay

6. Inrush current limit 7. e-Fuse/overload limit 8. Reverse voltaje 9. Power limit 10. Thermally controlled current foldba.

11. Current-Limiter with OV, UV, 12. and Reverse Protection 13. Current-Limiter with OV, UV, and Reverse Protection 14. Power Good o PWR_OK 15. OCP (Over-Current Protection) o Protección Contra Sobrecorriente 16. OVP (Over Voltage Protection) o Protección contra Sobretensión 17. UVP (Under Voltage Protection) o Protección contra Baja tensión 18. OPP (Over Power Protection) o Protección contra Sobrecarga 19. OTP (Over Temperature Protection) o Protección contra Temperatura Elevada 20. SCP (Short Circuit Protection) o Protección contra Cortocircuito

Figure 26. Turn-On and Turn-Off Characteristics of BJT

Insulated-Gate Bipolar Transistor (IGBT) IGBT combines the physics of both BJT and power MOSFET to gain the advantages of both worlds. It is controlled by the gate voltage. It has the high input impedance like a power MOSFET and has low on-state power loss as in case of BJT. There is no even secondary breakdown and not have long switching time as in case of BJT. It has better conduction characteristics as compared to MOSFET due to bipolar nature. It has no body diode as in case of MOSFET but this can be seen as an advantage to use external fast recovery diode for specific applications. They are replacing the MOSFET for most of the high voltage applications with less conduction losses. Its physical cross-sectional structural diagram and equivalent circuit diagram is presented in Fig. 27 to Fig. 29. It has three terminals called collector, emitter and gate.

Figure 27. IGBT Structure View

There is a p+ substrate which is not present in the MOSFET and responsible for the minority carrier injection into the n-region. Gain of NPN terminal is reduced due to wide epitaxial base and n+ buffer layer.

There are two structures of IGBTs based on doping of buffer layer:

a) Punch-through IGBT: Heavily doped n buffer layer ➔ less switching time b) Non-Punch-through IGBT: Lightly doped n buffer layer ➔ greater carrier lifetime ➔ increased conductivity of drift region ➔ reduced onstate voltage drop (Note: ➔ means implies)

Figure 28. Equivalent Circuit for IGBT

Figure 29. Simplified Equivalent Circuit for IGBT

Figure 30. Circuit Diagram for IGBT

Based on this circuit diagram given in Fig.30, forward characteristics and transfer characteristics are obtained which are given in Fig.31 and Fig.32. Its switching characteristic is also shown in Fig. 33.

Figure 31. Forward Characteristics for IGBT

Figure 32. Transfer Characteristics of IGBT

Figure 33. Turn-On and Turn-Off Characteristics of IGBT

(Note: Tdn : delay time ; Tr: rise time ; Tdf : delay time ; Tf1: initial fall time ; Tf2: final fall time)

Thyristors (SCR, GTO, MCT) Thyristors are the family of solid-state devices extensively used in power electronics circuitry such as SCR (silicon-controlled rectifier), DIAC

(diode on AC), TRIAC (triode on AC), GTO (gate turn-off thyristors), MCT (MOS-controlled thyristor), RCT, PUT, UJT, LASCR, LASCS, SIT, SITh, SIS, SBS, SUS, SBS and etc. SCR is the oldest member and the head of this family; and usually referred with the name “thyristor”.

They are operated as bistable switches that are either working in nonconducting or conducting state. Traditional thyristors are designed without gate-controlled turn-off capability in which the thyristor can come from conducting state to non-conducting state when only anode current falls below the holding current. While GTO is a type of thyristor that has a gate-controlled turn-off capability.

SCR SCR usually has three terminals and four layers of alternating p and ntype materials as shown in Fig. 34. The structure of the thyristor can be split into two sections: npn and pnp transistors for simple analysis purposes as shown in Fig. 36. It has three terminals named as cathode, anode and gate.

Figure 34. Structural View of Thyristor

N-base is a high-resistivity region and its thickness is directly dependent on the forward blocking rating of the thyristor. But more width of the n-

base indicates a slow response time for switching. Symbol of thyristor is given in Fig. 35.

Figure 35. Schematic Symbol of Thyristor

Figure 36. Two-Transistor Model of a Thyristor (A-Anode, G-Gate and K-Cathode)

Planar construction is used for low-power SCRs. In this type of construction, all the junctions are diffused.

For high power, mesa construction is used where the inner layer is diffused and the two outer layers are alloyed on it.

The static characteristic obtained from the circuit given in Fig. 37 is drawn in Fig. 38. It works under three modes: forward conducting mode, forward blocking mode and reverse blocking mode.

The minimum anode current that causes the device to stay at forward conduction mode as it switch from forward blocking mode is called the latching current. If the SCR is already conducting and the anode current is reduced from forward conducting mode to forward blocking mode, the

minimum value of anode current to remain at the forward conducting mode is known as the holding current.

Figure 37. Basic Circuit for Getting Voltage and Current Characteristics of Thyristor

Figure 38. Static Characteristics Curve of SCR Switching characteristics of SCR are shown in Fig. 39. Note that it can’t be turned off with the gate. This is due to positive feedback or a regenerative feedback effect.

Figure 39. Turn-On and Turn-Off Characteristics of SCR

GTO (Gate Turn-off Thyristor) GTO can be turned on with the positive gate current pulse and turned off with the negative gate current pulse. Its capability to turn off is due to the diversion of PNP collector current by the gate and thus breaking the regenerative feedback effect.

Actually the design of GTO is made in such a way that the pnp current gain of GTO is reduced. Highly doped n spots in the anode p layer form a shorted emitter effect and ultimately decreases the current gain of GTO for lower current regeneration and also the reverse voltage blocking capability. This reduction in reverse blocking capability can be improved by diffusing gold but this reduces the carrier lifetime. Moreover, it requires a special protection as shown in Fig. 43.

Fig. 40 shows the four Si layers and the three junctions of GTO and Fig. 41 shows its practical form. The symbol for GTO is shown in Fig.42.

Figure 40. Four Layers and Three Junctions of GTO

Figure 41. Practical Form of GTO

Figure 42. Symbol of GTO

Overall switching speed of GTO is faster than thyristor (SCR) but voltage drop of GTO is larger. The power range of GTO is better than BJT, IGBT or SCR. The static voltage current characteristics of GTO are similar to SCR except that the latching current of GTO is larger (about 2 A) as compared to SCR (around 100-500 mA).

The gate drive circuitry with switching characteristics is given in Fig. 43 and Fig. 44.

Figure 43. Gate Drive Circuit for GTO

Figure 44. Turn-On and Turn-Off Characteristics of GTO

MCT (MOS-Controlled Thyristor) IGBT is an improvement over a BJT using a MOSFET to switch on or switch off the anode current. Similarly, MCT is an improvement over a thyristor with a pair of MOSFETs to switch the current. There are several devices in the MCT’s family but the p-channel is commonly discussed.

Its schematic diagram and equivalent circuit is given in Fig. 45 and Fig. 46. Its symbol is given in Fig. 47.

Figure 45. Schematic Diagram of P-Type MCT

Figure 46. Equivalent Circuit for P-Type MCT

Figure 47. Symbol of P-Type MCT

Due to NPNP structure instead of PNPN, anode acts as a reference for gate. NPNP structure is represented by NPN transistor Q1 and a PNP transistor Q2 in the equivalent circuit.

The power required to switch it on or off is small with low switching losses due to its distributed structure across the entire surface of the device.

Delay time due to charge storage is also small. It also has a low on-state voltage drop.

When a p-type MCT is in the forward-blocking state, it can be switched on by applying a negative pulse to its gate (with respect to anode). While when an n-channel MCT is in the forward-blocking mode, it can be switched on with the positive gate pulse (with respect to cathode). It will remain on until the device current is reversed or a turn-off pulse is applied to the gate i.e. applying a positive gate pulse for p-type MCT with respect to anode.

This device can bear a high current and high didt didt capability. But just like any other devices, it needs to be protected against transient voltages and current spikes with the help of suitable snubbers. It is used in capacitor discharge applications, circuit breakers, AC-AC or ACDC conversion. It is an ideal replacement for GTO as it requires a much simpler gate drive and certainly more efficient.

SIMULA LABVIEW CONTROL POR FASE T1 tiempo de subida T2 tiempo de bajada Genera valor de b N=t1+t2 para el primer for Esto en fórmula nodo1 if(i<t1) b=(180/t1)*(i+1);

else b=b-(180/t2);

N=361 para el segundo for

Esto en fórmula nodo 2 y=156*sin(k); /*Rectificación*/ if(x<180) v=156*sin(k)*z/100; else v=156*sin(k-pi); /*Genera el pulso*/ if(b<x) z=100; else z=0;

if(x>180) z=0;

IRF840 MOSFET

CARACTERÍSTICAS DE SALIDA DE MOSFET CANAL N hola m1 2 1 0 0 IRF840 .lib PWRMOS.lib Vds 2 0 Vgs 1 0 .dc vds 0 15 0.5 vgs 0 6.5 0.5 .probe .end

TRANSCONDUCTANCIA -REEMPLAZO DE LA CARACTERÍSTICA DE ENTRADA PARA VDS=5 hola m1 2 1 0 0 irf840 .lib pwrmos.lib vds 2 0 5 vgs 1 0 .dc vgs 0 6 0.25 .probe .end

EN SATURACIÓN CON VGS=12 V PARA HALLAR VDS saturación hola m1 2 1 0 0 irf840 .lib pwrmos.lib vds 2 0 vgs 1 0 12 .dc vds 0 3 0.1 .probe .end

Para 180ma, vds=125mv con vgs=12v , para 2ª Vds=1.38v

TRABAJO MATEMÁTICO DEL CURSO-SIMULACIONES Monofásico, controlado, carga RL serie hola v1 1 0 sin(0 156 60 0 0 0) x1 1 2 3 2n1599 .lib THYRISTR.lib vg1 2 3 pulse(0 5 1.38m 1n 1n 10u 16.66m) r1 3 4 1k l1 4 0 50m ic=0 .probe .tran 41u 16.66m 0 41u uic .end

AhoraGráfico de v(3) y de AVG(v(3)) Ahora con el diodo de giro libre

hola v1 1 0 sin(0 156 60 0 0 0) x1 1 2 3 2n1599 .lib THYRISTR.lib vg1 2 3 pulse(0 5 1.38m 1n 1n 10u 16.66m) r1 3 4 1k l1 4 0 50m ic=0 .probe .tran 41u 16.66m 0 41u uic

d1 0 3 d1n4004 .lib diode.lib .end

I(r1), termina en 181,66° con disparo de 30°

Ahora con disparo em 90° hola v1 1 0 sin(0 156 60 0 0 0) x1 1 2 3 2n1599 .lib THYRISTR.lib vg1 2 3 pulse(0 5 4.16m1n 1n 10u 16.66m) r1 3 4 1k l1 4 0 50m ic=0 .probe

.tran 41u 16.66m 0 41u uic d1 0 3 d1n4004 .lib diode.lib .end

CARGA RLE, MONOFÁSICO, CONTROLADO, MEDIA ONDA DISCONTINUA

hola v1 1 0 sin(0 156 60 0 0 0) x1 1 2 3 2n1599 .lib THYRISTR.lib vg1 2 3 pulse(0 5 3.4m 1n 1n 20u 16.66m) r1 3 4 100 l1 4 5 100mh ic=0 v2 5 0 40 .probe .tran 41u 16.66m 0 41u uic d1 0 3 d1n4004 .lib diode.lib .end

El diodo giro libre está en paralelo con la RL, no con la fuente V2

RECTIFICADOR DE ONDA COMPLETA MONOFÁSICO, CARGA RL, CONDUCCIÓN DISCONTINUA O CONTINUA, DEPENDIENDO DE LA CARGA Y EL ÁNGULO DE DISPARO

Td=tiempo de disparo Ѳd=ángulo de disparo en radianes=wtd=6.28*f*t=377td f=frecuencia=60Hz 8.33m------180° x--------------- Ѳd°

CONDUCCIÓN DISCONTINUA, MONOFÁSICO ONDA COMPLETA, CONTROLADA, CARGA RL Ѳf°= ángulo de finalización de la corriente Ángulo de disparo 60°

DISCONTINUA

hola v1 1 2 sin(0 156 60 0 0 0) d1 0 1 d1n4004 d2 0 2 d1n4004 d3 0 3 d1n4004 .lib diode.lib

x1 1 5 3 2n1599 x2 2 6 3 2n1599 .lib THYRISTR.lib r1 3 4 100 l1 4 0 100m ic=0 vg1 5 3 pulse(0 5 2.08m 1n 1n 10u 16.66m) vg2 6 3 pulse(0 5 10.4m 1n 1n 10u 16.66m) .probe .tran 41u 16.66m 0 41u uic .end

FULL MONOFÁSICA NO CONTROLADA CARGA RL CONDUCCIÓ CONTINUA

hola v1 1 2 sin(0 156 60 0 0 0) d1 0 1 d1n4004 d2 0 2 d1n4004 d3 1 3 d1n4004 d4 2 3 d1n4004 .lib diode.lib r1 3 4 100 l1 4 0 100mh ic=352.7m .probe .tran 41u 16.66m 0 41u uic .end

Usamos un tiempo largo de simulación para encontrar el valor inicial de la corriente en la bobina en estado estable

MONOFÁSICA CONTROLADA, ONDA COMPLETA, CARGA RL, CONTINUA ѳd<=ꬿ=tan-1(wl/r)

hola v1 1 2 sin(0 156 60 0 0 0) d1 0 1 d1n4004 d2 0 2 d1n4004 d3 0 3 d1n4004 .lib diode.lib x1 1 5 3 2n1599 x2 2 6 3 2n1599 .lib THYRISTR.lib r1 3 4 100 l1 4 0 100m ic=0 vg1 5 3 pulse(0 5 0.46m 1n 1n 10u 16.66m) vg2 6 3 pulse(0 5 8.79m 1n 1n 10u 16.66m) .probe .tran 100u 50m 0 100u uic .end

AHORA LE PONEMOS LA CONDICIÓN INICIAL A LA CORRIENTE, 316.88m hola v1 1 2 sin(0 156 60 0 0 0) d1 0 1 d1n4004 d2 0 2 d1n4004 d3 0 3 d1n4004 .lib diode.lib x1 1 5 3 2n1599 x2 2 6 3 2n1599 .lib THYRISTR.lib

r1 3 4 100 l1 4 0 100m ic=316.68m vg1 5 3 pulse(0 5 0.46m 1n 1n 10u 16.66m) vg2 6 3 pulse(0 5 8.79m 1n 1n 10u 16.66m) .probe .tran 41u 16.66m 0 41u uic .end

Cambiamos L

hola v1 1 2 sin(0 156 60 0 0 0) d1 0 1 d1n4004 d2 0 2 d1n4004 d3 0 3 d1n4004 .lib diode.lib x1 1 5 3 2n1599 x2 2 6 3 2n1599 .lib THYRISTR.lib r1 3 4 100 l1 4 0 1h ic=890m vg1 5 3 pulse(0 5 0.26m 1n 1n 10u 16.66m) vg2 6 3 pulse(0 5 8.59m 1n 1n 10u 16.66m) .probe .tran 100u 50m 0 100u uic .end

CARACATERÍSTICAS DEL IRF740, 10 A 400V, 0.48Ω, TO220, GDS

Bootstrapping es una palabra inglesa que se utiliza en el mundo anglosajón para expresar la idea de “iniciar un negocio con pocos recursos” o “emprender únicamente con los medios que hay a tu alcance”.

Bootstrapping (electrónica): un tipo de retroalimentación positiva en un circuito analógico En el campo de la electrónica, un circuito bootstrap es aquel en el que parte de la salida de una etapa del amplificador se aplica a la entrada, para alterar la impedancia de entrada del amplificador

Bootstrapping en etapa de salida de audio.

Bootstrapping es un término de dudosa traducción. Se lo utiliza con distintos significados en economía, informática y electrónica. En esta última no siempre para una misma función.

Los diseñadores buscan sacar el mayor rendimiento de un amplificador. Para eso la salida pico a pico debería ser igual a la tensión de fuente. Sin embargo, esto es un máximo teórico imposible de alcanzar.

En el primer circuito vemos un amplificador básico de salida complementaria. Los valores de R2, R3 y R4 son tales que en los emisores del par de salida la tensión en reposo es la mitad de la tensión de la fuente. El resistor R4 es la carga del transistor TDriver y los dos diodos están para evitar la distorsión de cruce en el par de salida.

El circuito funciona de forma que el transistor de salida TS1 conduce en el hemiciclo positivo de la señal presente en las bases del par de salida y el TS2 en el hemiciclo negativo. El capacitor C2 recibe corriente desde la fuente a través de TS1 y la cede a masa a través del transistor TS2.

Si pretendemos aprovechar al máximo la excursión pico a pico, suceden cosas que nos obligan a hacer ajustes. Supongamos, primero, que queremos llegar al máximo en el hemiciclo positivo de la señal de entrada. El pico positivo en la señal de entrada corresponde al valor mínimo de tensión con respecto a masa en la salida.

Quisiéramos que fuera 0 V, pero, aunque la tensión de la base de TS2 fuera nula, el emisor de TS2 estaría 0,6 V más alto. De forma que el

mínimo de la señal llegará a ese valor, a lo sumo.

Supongamos que la tensión de la fuente es de 12 V. De esta forma, el pico respecto de 6 V será de - 5,4 V. Ahora pensemos en la otra mitad de la señal, la que amplifica TS1. Si TDriver llega al corte, en el extremo de R4 que conecta el colector de TDriver y la base de TS1 podemos tener 12 V (despreciamos los diodos). Así, el emisor de TS1 debería estar 0,6 V por debajo de esa tensión. Pero, el transistor de salida TS1 no tiene ganancia de corriente infinita; todo lo contrario, es baja, bien finita. Como no podía ser de otra manera, TS1 toma una corriente de base importante, que provoca una caída de tensión significativa en R4. Si, por ejemplo, en esas condiciones hay una tensión de 10,5 V en la base de TS1, en su emisor habrá 9,9 V. O sea, respecto de 6 V tendremos 3,9 V. A partir de ahí, si le pedimos más, habrá recorte. El recorte será a partir de + 3,9 V y de - 5,4 V respecto de 6V. Si le exigimos el máximo al amplificador, el recorte no será simétrico.

Esto podría evitarse conectando R4 a una fuente de 15 V, pero encarece el equipo. De forma que los ingenieros se las arreglaron para obtener esa tensión de las que están presentes en el circuito.

Si reducimos R4 a la mitad e insertamos otra resistencia R5 = R4 en serie, conectando un capacitor C4 entre ellas y el punto de salida, éste se cargará con 3 V. Si este capacitor es de un valor tal que no se descargue ante las variaciones rápidas del punto de salida, siempre su borne positivo suministrará sobre R4 una tensión 3 V por encima de la que hay en la salida. De esta forma permite que la base de Ts1 tenga

la tensión necesaria para que el recorte sea parejo.

Esto no es otra cosa que una realimentación positiva, que hace lo contrario que una realimentación negativa. Producirá inestabilidad, aumento de ruido y distorsión; especialmente fuera del régimen senoidal.

Vemos el circuito modificado en la siguiente imagen.

Pero el ahorro no es completo, tenemos un capacitor más que antes.

Para evitar C4, se conecta el parlante o altavoz a la fuente, C2 es invertido, para respetar la polaridad, y R4 se alimenta desde la unión de C2 y el altavoz.

El siguiente circuito nos muestra valores prácticos para una salida de 250 mW sobre 8 ohmnios, como podría ser el amplificador de audio en una radio. El driver toma la alimentación de la salida, con el propósito de aumentar la estabilidad.

Este circuito tiene dos inconvenientes inmediatos: * Circula corriente continua por el parlante. * Si se desconecta el parlante cambia la polarización del par de salida.

Actualmente los equipos de estado sólido de calidad están acoplados en continua y son diferenciales de entrada a salida. Tienen al menos

dos fuentes de alimentación y la menor realimentación negativa posible.

Transistor Bootstrapping Circuit Using BJT Bootstrapping is a technique used in the design of transistor amplifier circuits to increase the input impedance and thereby reduce the loading effects on the input source. It typically involves the use of a bootstrap capacitor, which provides positive feedback of ac signals to the base junction of a transistor in an emitter follower circuit. Application of this feedback increases the effective value of the base resistance. The amount of increase is a factor determined by the voltage gain of the circuit.

A bipolar junction transistor (BJT) has an inherently low input impedance of typically 1 Ω to 50 kΩ, and this reduces further after the addition of biasing transistors such as a potential divider network connected to the base junction of a transistor. Whilst it improves dc stability, these resistors have the effect of shunting the input thereby reducing the input resistance. These types of circuits usually pose a problem where high input impedance is required, and therefore bootstrapping is a technique that helps increase the input impedance.

A high impedance input is very useful for example in radio circuits where the base junction of a transistor connects to a ferrite coil to amplify the radio signal. Usually the amplifier stage is required to minimise the loading on the coil for better amplification. It is also useful in

instrumentation where you need to take accurate measurements using probes such as bio-probes.

A common emitter follower circuit such as the one shown above will achieve a one hundred-fold increase in the effective value of Rb, when the voltage gain is 0.99. For a further increase in impedance, usually a two stage direct-coupled circuit is utilised.

Related Articles Transistor Bootstrapping Circuit Transistor Bootstrapping Circuit Using FET

Using

BJT

SIIMULACIÓN SINCRONISMO DE RED CON ZENER NO FUNCIONA

hola v1 1 2 sin(0 156 60 0 0 0) d1 1 3 d1n4004 d2 2 3 d1n4004 d3 0 1 d1n4004 d4 0 2 d1n4004 d5 0 4 D1N4742 r2 3 0 10k

.lib diode.lib r1 3 4 10k .probe .tran 41u 16.66m 0 41u .end

SEÑAL OBTENIDA DE SINCRONISMO DE RED

Se utiliza para disparar un LM555 como Timer y luego invertir para disparar tiristores

D1N4742 .model D1N4742-X

D(

Is=1.773f Rs=2.06 Ikf=0 N=1 Xti=3 Eg=1.11 Cjo=100p M=.3894 Vj=.75 Fc=.5 Isr=1.799n Nr=2 Bv=12 .model D1N4742-X Is=1.773f Rs=2.06

D(

Ikf=0 N=1 Xti=3 Eg=1.11 Cjo=100p M=.3894 Vj=.75 Fc=.5 Isr=1.799n Nr=2 Bv=12)

Detector de cruce cero con optoacoplador

RECTIFICADOR DE MEDIA ONDA WILAEBA ELECTRÓNICA

RECTIFICADORCONTROLADO DE ONDA COMPLETETA 4 TIRISTORES EN PUENTE CARGA R

optoacopladores pc817 - 2 optotriac moc3021 - 2 diodos 1n4007 - 1 CI lm324 (contiene 4 amplificadores operacionales) - 2 transistores 2n2222a - 4 SCRs tyn612

RECTIFICAADOR TIFASICO DE MEDIA ONDA CONTROLADO

R

ECTIFICADOR TRIFÁSICO DE ONDA COMPLETA

CAPÍTULO V. TODAS LAS PROTECCIONES ELECTRÓNICAS EN MEDIA Y BAJA TENSIÓN. CP: CURRENT PROTECTION Fuses Fast Blow Electronic Fuse or Reseteable Fuses . PTC thermistor as a resettable fuse.

Electrical, Specialty Fuses TC Resettable Fuses Thermal Cutoffs (Thermal Fuses) Fuseholders Relais, Relés o relevadores Circuit Breakers and stoers Gas Discharge Tube Arresters (GDT) Inrush Current Limiters (ICL)( Ground Fault Circuit Interrupter (GFCI)( Circuit breaker Electrostatic discharge MCB. Mini Circuit Breaker ·

Acronyms, Acrónimos

BPCP

Bypass Current Protection

FBCLP

Fold Back Current Protection

SCP

Short Circuit Protection

SDCP

Shut Down Current Protection

FFCLP

Fold Forward Current Limiting Protection

OLP

Over Load Protection

NWP

Net Working Protection

OCLP

Over Current Limiting Protection

FEEDFORWARD

Progresive Current Protection

RCP

Residual Current Protection

HICCUP

Improvement hiccup mode over-current protection-Rearme temporizado de corriente

ATSP

Automatic Transfer Switch Protection

ICP

Inrush Current Protection

WP

Wire Protection

CLP

Constant Limiting Protection

PLSP

Power Line Surge Protection

ILCP

Level interlocking current protection. enclavamiento

V:P :OLTAGE PROTECTION OVP

Over Voltage Protection-circuito de palanca-

OVP

Crowbar -voltge-variable reference or programable voltage reference

OVP

MOV

UVP

Under Voltage Protection

UVP

LOMP- Loss Of Main protection

SSICs

Surge Suppression ICs

SSICs

Surge Suppression ICs

TVS -

Diodes

TVS -

Mixed Technology(941 Items)

TVS -

Surge Protection Devices (SPDs

TVS -

Thyristors

TVS -

Varistors, MOVs

Oters item protections Thermal Cutoffs (Thermal Fuses Protective Relaying Devices Breaker Failure Relay Protection Industrial and Commercial Protection Transformer Differential Relay Protection Sobre Carga Corto Circuito Caída de Tensión Elevación de Tensión 8 Inversión del Flujo de Potencia Variación de Frecuencia Protección de respaldo Falla de fase}Falla datierra contra sobrecargas y cortocircuitos. Selectividad de las protecciones de sobreintensidad.

CIRCUIT PROTECTION DEVICES Circuit Breakers Fuse Holders, Clips & Blocks Resettable Fuses (PPTC) - Polyswitch & Polyfuse Electronic Fuses Gas Discharge Tubes (GDTs) Thyristor Surge Protection Devices (TSPD) ESD Protection & Diode Arrays Protection Thyristors hot circuit protection devices Fuses, Circuit breaker ,Surge protection ICs Electrostatic discharge (ESD) protection devices The supercapacitor auto-balancing (SAB) MOSFETs from Advanced Linear Devices (ALD) automatically balance the PTC thermistors provide stability and reliability in high-voltage applications and are optimized for inrush current Protection thyristor for data lines

The Littelfuse thyristor is designed for the protection of lowvoltage signal-line applications. (Image: Littelfuse) The P0080S4BLRP features minimum operating surge peak current capability. variety of temperature-current combinations. (Image: Bourns) . Surge protection for mobile devices . PTC thermistor as a resettable fuse voltage range for unidirectional for bidirectional devices.

Transient overvoltages, overvoltaces in power lines Transient suppressors Crowbar circuits Varistors Zeners diodes Avalancha diodes Protecciones con circuitos de palanca Contactor, limitación de corriente, interruptor automático Electronic Circuit Protections 1. Descargas inductivas 2. Current Limiting Circuit Breaker 3. Redundancia

4.

5.

6.

7.

8.

Fuses and Thermo-magnetic circuit breakers A transient voltage suppressor (TVS) gas discharge tube (GDT) METAL OXIDE VARISTOR MOV VOLTAGE (OVP, Crowbar,Clamping, . Caída de Tensión,Elevación de Tensión, Protección de transformadores.

Por ovp, Variación de Frecuencia, UVP, Loss,Ffrout, Transient, UPS, APS, Bypass, Consumo, FP, Microcortes de red, Cortes de red, Suspensiones, Restableciniento, Forma de onda, Harmónicos indeseables, Descargas atmosféricas, Interferencias,Fallas en el ciclo de servicio, fallas de descargas de líneas de potencia, peaking detector circuits….

9. Lightning Protection 10. Protection Thyristor Devices: TVS Diodes, ESD Protection-MLV (multi-layer varistor) 11. CURRENT (OCP, SCP, OLP, Foldback,Ffeedforwaed, sco, Bypass, Limitin current, Starting, Constant current, Shutdown, Start circuits, remote on-off, peaking detector circuits 12. Inversión del Flujo de Potencia

,WIRE( errores de acometida, fallas entre lineas, fallas a tierra, tierra, interferencias, uso inapropiado del calibre de conductores, errores en cableado estructturado, temperatura, blindaje, cableado a tierra, …….

PROTECCIONES ELECTRÓNICAS ASOCIADAS A LOS DISPOSITIVOS DE ESTADO SÁÓLIDO, temeratura, salida de los ratings recomendados, malas conexiones en tableros…….

PROTECIIONES PARA SERES UMANOS Y ANIMALES( Fallas en el seguimiento de protocolos,Hherramientas inapropiadas, Comportamiento frente a accidentes eléctricos……..

PARA FUENTES DC LINEALES Power Good o PWR_OK OCP (Over-Current Protection) o Protección Contra Sobrecorriente OVP (Over Voltage Protection) o Protección contra Sobretensión UVP (Under Voltage Protection) o Protección contra Baja tensión OPP (Over Power Protection) o Protección contra Sobrecarga OTP (Over Temperature Protection) o Protección contra Temperatura Elevada SCP (Short Circuit Protection) o Protección contra Cortocircuito

circuito de monitorización. medir la corriente en las salidas

he Crowbar

The Clamp

Fast TTL Power Supply protection using SCR You may know that TTL requires a constant 5V power supply. Even a little over-voltage. It also can destroy TTL chips very rapidly.

And, it doesn’t happen often. When using the 7805 regulators.

The use of a normal fuse is a good and a save money choice.

The TTL safe from too high. Because voltage is low almost is 0V. While Fust will blow very fast.

So, IC-78xx series is saved from over-current too.

As it happens, this is the most common type of stabilizer failure, so that the protection is necessary for any TTL circuit of any significance.

Some use CMOS in digital circuits. Because they use wide voltage power supply 3V to 16V. But most circuits use more TTL inside and still use 5V power supply.

So, the power protection circuit has necessary for the Regulator and others.

Fast TTL Power Supply protection using SCR How it works In the circuit above. If the voltage at the output terminals rises above 6.2V. The Zener diode conducts charging capacitor C4.

This voltage will trigger the silicon-controlled rectifier (SCR) turn-on.

It will quickly short circuit output to ground very fast. The supply rails blow the fuse immediately.

How it works When entering the input voltage to the circuit. Both transistors Q1 and Q2 are in a bistable multivibrator circuit.

First, Q2 will start. But Q1 stops. Because C1 connected to the base of Q1. It is charging. So, the collector voltage of Q1, to bias the base pin of Q3 through R4. That cause Q3 conducts.

And Q4 will conduct too.

So, there is a voltage input to pin 1 of IC1. It causes the output voltage any load.

However, if there is a short circuit. D1 works like connecting to the ground.

That makes the Q2 base have voltage is about 0.3V. Then, Q2 stop the working of the bistable circuit.

They change the operating to cause Q1 runs. So, Q3, Q4, and 78xx IC stop working. So, no the output voltage.

If you want IC works again. You need to press S1 switch. To reset the circuit to return the working.

Or the bistable multivibrator to work and control Q4 work again too.

Parts lists

Q1-Q3: BC547 NPN transistors Q4: TIP125 Darlington transistor D1: BAT85 Diode IC1: 7800 Regulator series S1: Pushbutton switch 0.25 Resistors, tolerance 5% R1, R5: 4.7K

R2, R3: 22K R4: 100K R5: 4.7K R6: 3.3K C1: 0.1uF 50V Ceramic You may also like these:

LM350 adjustable voltage Regulator Variable power supply using LM317 Fixed voltage regulator working principle LM317 Low dropout regulator 5V 2A using TIP41

0-12v variable power supply at 3ª

http://www.industrial-electronics.com/glossary.html

A |B|C|D|E|F|G|H|I|J|K|L|M|N|O|P|Q|R|S|T|U|V| W|Y|Z ac-coupled amplifier An amplifier with a capacitive-coupled input to filter out the dc component of the input signal. ac input circuit A circuit on a PLC I/O module that converts ac signals from a pilot switch voltage level to backplane logic-level dc signals. ac input module An I/O module for a PLC or other controller that contains circuits that convert ac signals to backplane logic-level dc signals. ac output circuit A circuit on a PLC UO module that converts backplane logic-level dc signals to field-level ac voltage. ac output module A PLC I/O module that contains circuits that convert backplane logic-level dc signals to field-level ac voltage. across-the-line starter A motor starter that allows full voltage to be applied to a motor when the contacts of the starter are closed. actuator A device that converts an electrical signal into motion. address A character string that uniquely identifies a memory location in a PLC, computer, or other programmable system. alternating current Current that changes from a positive level to a negative level periodically. Example: a 60 Hz alternating current change from positive to negative 60 times per second.

ambient temperature The temperature of the medium (air, water, earth).

American wire gauge (AWG) A standard used for identifying the size of electrical conductors. Gauge numbers have an inverse relationship to size: larger numbers have a smaller cross-sectional area. analog circuit A circuit in which the signal can vary continuously between specified limits from some minimum to some maximum. analog input module A PLC UO module that contains circuits that convert analog dc input signals to digital values. analog output module A PLC UO module that contains circuits that convert digital values to an analog dc signal. analog-to-digital conversion (A/D conversion) A circuit that converts analog values to digital (numerical) values. angstrom Unit of measure for the wavelength of light. 1 ångström (Å) = 10–10 meters = 0.1 nm ANSI American National Standards Institute. An organization in the United States that develops standards for industry. armature The moving part of a magnetic circuit, such as the rotating part of a motor or a generator or the movable iron part of a relay.

ASCII American Standard Code for Information Interchange. It is a seven-bit code with an optional parity bit used to represent alphanumerics, punctuation marks, and control-code characters.

P

A|B|C|D|E|F|G|H|I|J|K|L|M|N|O|P |Q|R|S|T|U|V|W|Y|Z photoelectric control An electronic device that recognizes changes in light intensity and converts these changes into a change in electrical output state. pilot circuit The portion of a control circuit that carries the controlling signal for a device which, in turn, controls the primary current.

PLC controller A programmable logic controller such as an AllenBradley programmable controller. plugging Reversing either line voltage polarity or phase sequence on a motor so that the motor develops a counter-torque that exerts a retarding force to brake the motor. position loop A feedback control loop that uses an encoder or resolver as a feedback device to indicate mechanical position.

position transducer An electronic device (e.g., encoder or resolver) that measures incremental or absolute position and converts this measurement into a feedback signal. positive feedback A feedback signal from the output that is added to the input signal. power Work done per unit of time. Measured in horsepower or watts: 1 hp = 33,000 ft-lb/min = 746 W. power factor A measurement of the time phase difference between the voltage and current in an ac circuit. It is represented by the cosine of the angle of this phase difference. Power factor is the ratio of real power (in watts) to apparent power (in volt-amperes). power supply A device that converts available power to a form that a system can use: usually converts ac power to dc power. pressure switch A switch that is activated at a specific pressure. printed circuit board A board (card) made up of a nonconductive layer sandwiched by conductive layers that are etched to form circuit connections between connection points where components can he mounted. processor The decision-making and data storage sections of a programmable controller or computer. programmable controller A solid-state control system that continually scans its user program. The controller has a user-programmable memory for storage of instructions to implement specific functions such

as I/O control, logic, timing, counting, report generation, communications, arithmetic, and data file manipulation. A controller consists of a central processor unit, input/output interface, and memory. A controller is designed as an industrial control system. The program is generally in the form of ladder logic. program mode On a programmable controller, a mode in which the ladder logic is not executed and all outputs are held off.

PROM Programmable read-only memory. A type of ROM that requires an electrical operation to store data. In use, bits or words are read on demand but never changed. proportional band The range of values through which the output is proportional to the change of the input variable. proportional, integral, derivative control (PID) A method of control that uses proportional (gain), integral (reset), and derivative (rate) to provide control of the response rate of the output signal.

Proportional control causes an output signal to change as a direct ratio of the error signal variation.

Integral control causes an output signal to change as a function of the integral of the error signal over the time duration.

Derivative control causes an output signal to change as a function of the rate of change of the error signal. protocol A set of conventions governing the format and timing of data between communications devices.

proximity switch/sensor A switch/sensor that is actuated when an actuating device is moved near it, without physical contact.

PWM Pulse-width modulation. A technique used to eliminate or reduce unwanted harmonic frequency when inverting dc voltage to sine wave ac.

A|B|C|D|E|F|G|H|I|J|K|L|M|N|O|P |Q|R|S|T|U|V|W|Y|Z photoelectric control An electronic device that recognizes changes in light intensity and converts these changes into a change in electrical output state. pilot circuit The portion of a control circuit that carries the controlling signal for a device which, in turn, controls the primary current.

PLC controller A programmable logic controller such as an AllenBradley programmable controller. plugging Reversing either line voltage polarity or phase sequence on a motor so that the motor develops a counter-torque that exerts a retarding force to brake the motor. position loop A feedback control loop that uses an encoder or resolver as a feedback device to indicate mechanical position.

position transducer An electronic device (e.g., encoder or resolver) that measures incremental or absolute position and converts this measurement into a feedback signal. positive feedback A feedback signal from the output that is added to the input signal. power Work done per unit of time. Measured in horsepower or watts: 1 hp = 33,000 ft-lb/min = 746 W. power factor A measurement of the time phase difference between the voltage and current in an ac circuit. It is represented by the cosine of the angle of this phase difference. Power factor is the ratio of real power (in watts) to apparent power (in volt-amperes). power supply A device that converts available power to a form that a system can use: usually converts ac power to dc power. pressure switch A switch that is activated at a specific pressure. printed circuit board A board (card) made up of a nonconductive layer sandwiched by conductive layers that are etched to form circuit connections between connection points where components can he mounted. processor The decision-making and data storage sections of a programmable controller or computer. programmable controller A solid-state control system that continually scans its user program. The controller has a user-programmable memory for storage of instructions to implement specific functions such

as I/O control, logic, timing, counting, report generation, communications, arithmetic, and data file manipulation. A controller consists of a central processor unit, input/output interface, and memory. A controller is designed as an industrial control system. The program is generally in the form of ladder logic. program mode On a programmable controller, a mode in which the ladder logic is not executed and all outputs are held off.

PROM Programmable read-only memory. A type of ROM that requires an electrical operation to store data. In use, bits or words are read on demand but never changed. proportional band The range of values through which the output is proportional to the change of the input variable. proportional, integral, derivative control (PID) A method of control that uses proportional (gain), integral (reset), and derivative (rate) to provide control of the response rate of the output signal.

Proportional control causes an output signal to change as a direct ratio of the error signal variation.

Integral control causes an output signal to change as a function of the integral of the error signal over the time duration.

Derivative control causes an output signal to change as a function of the rate of change of the error signal. protocol A set of conventions governing the format and timing of data between communications devices.

proximity switch/sensor A switch/sensor that is actuated when an actuating device is moved near it, without physical contact.

PWM Pulse-width modulation. A technique used to eliminate or reduce unwanted harmonic frequency when inverting dc voltage to sine wave ac.

http://www.industrial-electronics.com/index2.htmlprotections Popular Electronics--Electronic Experimenter's Handbook (1980a) Computer Buying Tips Getting Your System Up and Running Perf Board Wiring Techniques for Experimenters The Oscilloscope Graphic Artist A BCB Loop Antenna for DX'ing The "Bucket Brigade" Audio Delay Build This Low-Cost Capacitance Meter Build A Digital Camera Shutter Timer Automatic Diode Checker Measure RPM of Rotating Elements with the IC Photo Tachometer

Build the Audio Detective Power-Failure Alarm Designing Optimum-Q and Small Inductors A Strobe Flasher for Night Cycling A Power Nomograph IC Multiplex Decoder Improves FM Performance Quiz of Audio Basics Build a Direct Reading Logic Probe Talk Over A Sunbeam With A "Photophone" A "Junk-Box" 5-Volt Power Supply Build the "Light Genie" Build A State-Of-The-Art Battery Charge Monitor

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Four low-cost projects Build an In-circuit Transistor Tester for $15 How to Add Triggered Sweep To An Oscilloscope Designing Circuits for Worst-Case Operation Build The "Super Marker" Precision References for Current & Voltage Build a Pink Noise Generator For Audio Testing

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OVP

CROWBARD PROTECTION

S

USANDO ZENER PROGRAMABLE Y TRIAC

OVER CURRENT Cuando la corriente de carga se eleva por encima de un valor predeterminado, el pequeño voltaje desarrollado en R5 será suficiente (a aproximadamente 0,7 v) para encender Tr3. Como Tr3 está conectado a través de la unión base / emisor del transistor de control principal Tr1, la acción de encender Tr3 reducirá el voltaje base / emisor de Tr1 en una cantidad que depende de la cantidad de exceso de corriente. No se permitirá que la corriente de salida aumente por encima de una cantidad predeterminada, incluso si ocurre un cortocircuito completo en los terminales de salida. En este caso, el voltaje base / emisor de Tr1 se reducirá prácticamente a cero voltios, lo que evitará que Tr1 conduzca. En estas condiciones, el voltaje de

salida caerá a cero mientras persista la condición de exceso de corriente, pero el suministro no sufrirá daños.

Proteccion al sobrevoltaje.

Cuando se utilizan suministros regulados, el voltaje de entrada de CC al regulador suele ser considerablemente más alto que el voltaje de

salida requerido. Por lo tanto, si ocurre una falla en la fuente de alimentación, es posible que el voltaje de salida regulado suba repentinamente a un nivel que pueda dañar otros componentes. Por esta razón, es común encontrar protección contra sobretensión incluida en suministros estabilizados. El circuito que se muestra en la figura 2.2.6 a veces se llama circuito de "palanca" porque cuando opera, coloca un cortocircuito completo a través de la salida, ¡un efecto similar a dejar caer una palanca de metal a través de los terminales de salida positivo y de tierra!

Operación del circuito de palanca.

En la figura 2.2.6, el diodo Zener D Z 2 tiene un voltaje de ruptura ligeramente menor que el valor máximo permitido para V OUT . El resto de V OUT se desarrolla en R6, VR2 y R7.

VR2 es un potenciómetro, por lo que se puede tomar un voltaje de la red de resistencias para polarizar correctamente el diodo D1. Este diodo tiene su cátodo mantenido a 0 V por R8, y VR2 se ajusta de modo que D1 esté fuera de conducción, es decir, su voltaje de ánodo es aproximadamente 0,5 V más alto que su voltaje de cátodo.

Ahora, si V OUT aumenta, el voltaje en R6, VR2 y R7 aumentará en la misma cantidad, ya que el voltaje en D Z 2 seguirá siendo el mismo.

Por lo tanto, habrá un aumento sustancial en el voltaje en el control deslizante R7, lo que hará que D1 conduzca, suministrando un pulso de corriente a la puerta del tiristor Th1, provocando que se "dispare" y conduzca fuertemente hasta que V OUT caiga prácticamente a 0v. Se

incluye R9 para limitar el flujo de corriente resultante a través del tiristor a un nivel seguro.

La gran corriente que fluye cuando Th1 se dispara ahora hará que el circuito limitador de corriente entre en funcionamiento como se describió anteriormente. Esto apagará de manera segura el suministro hasta que la sobrecorriente causada por Th1 haya desaparecido, lo que por supuesto sucederá tan pronto como V OUT llegue a 0 V, pero si el sobrevoltaje aún estuviera presente cuando Th1 se apague y V OUT aumente nuevamente, el circuito se volverá a disparar, haciendo que el voltaje en la carga alterne repetidamente entre su valor normal y cero; un síntoma inofensivo pero claro de un problema de sobretensión.

OVP CIRCUITOS DE PALANCA Con TL431 Zener Programabe

https://www.electronicdesign.com/power-management/powerprotection/whitepaper/21120596/protection-for-the-power-supply-andits-load

ELECTRONIC PROTECTIONS IN DC POWER SUPPLY 1. Begins with Overcurrent and Fuses 2. Undervoltage Lockout (UVLO) 3. Overvoltage Protection (OVP)

4. The Crowbar

5. The 5. Clamp

6. Residual Current Device (RCD) 7. Miniature Circuit Breaker (MCB)

8. TVS Diodes: A TVS, or transient voltage suppression diode

9. MOV/MLV Varistors: A varistor is a resistor that has a non-ohmic current; 10.

Zener diodes 11.

Resettable Fuses 12.

TVS diodes: Transient Voltage Suppressors (TVS) 13.

Thyristors: Thyristors, or TPSD (thyristor surge protection devices), 14. hermal Cutoffs: As opposed to most other circuit protection devices, thermal cutoffs 15.

Circuit breakers: Generally thought of as being in higher voltage systems, circuit breakers 16.

Protect against short-circuit using a resettable eFuse 17.

Use eFuse to fulfill IEC 62368-1 requirements 18.

Compare integrated OVP and OCP solutions vs. discrete solutions 19.

Protect against ESD short-pulse transients vs power supply transient and noise 20.

Design protection for USB power delivery 21. current surge and overvoltage protection 22.

OCP

1.

2.

TIMER CIRCUITS

TRIGGER CIRCUIT ELECTRONIC PROTECTION ENGENIERIN IEEE (CAP 6) LECTURAS COMPLEMENTARIAS (CAP 4, 5 6)

CP: CURRENT PROTECTION Fuses Fast Blow Electronic Fuse or Reseteable Fuses . PTC thermistor as a resettable fuse.

Electrical, Specialty Fuses TC Resettable Fuses Thermal Cutoffs (Thermal Fuses) Fuseholders Relais, Relés o relevadores Circuit Breakers and stoers Gas Discharge Tube Arresters (GDT) Inrush Current Limiters (ICL)( Ground Fault Circuit Interrupter (GFCI)( Circuit breaker Electrostatic discharge MCB. Mini Circuit Breaker ·

Acronyms, Acrónimos BPCP

Bypass Current Protection

FBCLP

Fold Back Current Protection

SCP

Short Circuit Protection

SDCP

Shut Down Current Protection

FFCLP

Fold Forward Current Limiting Protection

OLP

Over Load Protection

NWP

Net Working Protection

OCLP

Over Current Limiting Protection

FEEDFORWARD

Progresive Current Protection

RCP

Residual Current Protection

HICCUP

Improvement hiccup mode over-current protection-Rearme temporizado de corriente

ATSP

Automatic Transfer Switch Protection

ICP

Inrush Current Protection

WP

Wire Protection

CLP

Constant Limiting Protection

PLSP

Power Line Surge Protection

ILCP

Level interlocking current protection. enclavamiento

V:P :OLTAGE PROTECTION OVP

Over Voltage Protection-circuito de palanca-

OVP

Crowbar -voltge-variable reference or programable voltage reference

OVP

MOV

UVP

Under Voltage Protection

UVP

LOMP- Loss Of Main protection

SSICs

Surge Suppression ICs

SSICs

Surge Suppression ICs

TVS -

Diodes

TVS -

Mixed Technology(941 Items)

TVS -

Surge Protection Devices (SPDs

TVS -

Thyristors

TVS -

Varistors, MOVs

Oters item protections Thermal Cutoffs (Thermal Fuses Protective Relaying Devices Breaker Failure Relay Protection Industrial and Commercial Protection Transformer Differential Relay Protection Sobre Carga Corto Circuito Caída de Tensión Elevación de Tensión 8 Inversión del Flujo de Potencia Variación de Frecuencia Protección de respaldo Falla de fase}Falla datierra contra sobrecargas y cortocircuitos. Selectividad de las protecciones de sobreintensidad.

CIRCUIT PROTECTION DEVICES Circuit Breakers Fuse Holders, Clips & Blocks Resettable Fuses (PPTC) - Polyswitch & Polyfuse Electronic Fuses Gas Discharge Tubes (GDTs) Thyristor Surge Protection Devices (TSPD) ESD Protection & Diode Arrays Protection Thyristors hot circuit protection devices Fuses, Circuit breaker ,Surge protection ICs Electrostatic discharge (ESD) protection devices The supercapacitor auto-balancing (SAB) MOSFETs from Advanced Linear Devices (ALD) automatically balance the PTC thermistors provide stability and reliability in high-voltage applications and are optimized for inrush current

Protection thyristor for data lines The Littelfuse thyristor is designed for the protection of lowvoltage signal-line applications. (Image: Littelfuse) The P0080S4BLRP features minimum operating surge peak current capability. variety of temperature-current combinations. (Image: Bourns) . Surge protection for mobile devices . PTC thermistor as a resettable fuse voltage range for unidirectional for bidirectional devices.

Transient overvoltages, overvoltaces in power lines Transient suppressors Crowbar circuits Varistors Zeners diodes Avalancha diodes Protecciones con circuitos de palanca Contactor, limitación de corriente, interruptor automático

ALLELECTRONIC PROTECTIONS

OVP (Over Voltage Protection): A power supply feature which shuts down the unit, or suspends the output, when the voltage exceeds a preset level. It is generally activated when the voltage output exceeds 110% to 130%.

OPP (Over Power Protection): Prevents damage resulting from excessive power output. This is usually activated when the power to connected components reaches 130% to 150%.

OCP (Over Current Protection): Protects against the potential dangerous effects of pushing too much current into the PSU. This can cause the unit to overload or short-circuit, potentially creating a faulty current and damaging a power supply unit or connected components

such as the motherboard. It will be activated when the export current reaches 130% to 150%.

Ironically, this protection setting is often the main reason systems can be unstable. This is because sometimes the motherboard and the graphics card are pushed to the point that current needs set of this protection prematurely, as in the system is working properly, but the amount of current is outside the predetermined range of safety, ultimately triggering the OCP and shutting down the system.

SCP (Short Circuit Protection): Prevents the motherboard from burning due to high temperature output.

OTP (Over Temperature Protection): Shuts down the power supply when the internal temperature exceeds the maximum safe operating temperature.

BOP (Brown Out Protection): Prevents a power supply unit from being damaged due to a sudden drop in voltage from inconsistent electrical grids.

* OVP (OverVoltage Protection): Apaga la fuente en caso de sobre voltaje. * UVP (UnderVoltage Protection): Apaga la fuente en caso de bajo voltaje.

* OCP (OverCurrent Protection): Apaga la fuente si se demanda mas amperaje del permitido en un riel. * OPP/OLP (OverPower/OverLoad Protection): Apaga la fuente si se demanda mas potencia de la permitida. * OTP (OverTemperature Protection): Apaga la fuente si se excede cierta temperatura de funcionamiento. * SCP (ShortCircuit Protection): Apaga la fuente en caso de corto circuito.

Contents • • • • • • •

1. Introduction 2. Power Good 3. Under and Over Voltage Protections (UVP and OVP) 4. Over Current Protection (OCP) 5. Over Temperature Protection (OTP) 6. Other Protections 7. Comparison Between Monitoring Integrated Circuits

Power Good When we first turn on the power supply, voltages are not immediately available on the power supply outputs: they increase until reaching

their correct values. This increase happens is a fraction of a second (maximum of 20 ms or 0.02 s to be more exact).

There is also another reason for this signal to exist: the under voltage protection (UVP This signal is generated by the monitoring integrated circuit or by the PWM controller (in the case of power supplies based on the half-bridge topology).

Everything You Need to Know About Power Supply Protections

By Gabriel Torres - January 18, 2010 9535 SHARE Facebook Twitter Under and Over Voltage Protections (UVP and OVP) shut down the power supply in case any of these outputs are above (OVP) or under (UVP) a certain value, also called “trigger point.” These are the most basic protections available over voltage protection (OVP) but the under voltage protection (UVP) is optional.

The problem with these two protections is that they are usually configured at trigger points that are too far away from the output

nominal voltage. To give you a better idea, consider the over voltage protection trigger points required by the ATX12V specification:

Other Protections Over Power/Load Protection (OPP/OLP)

Over Power Protection (OPP) and Over Load Protection (OLP) are two different names for the same thing. This is an optional protection that shuts down the power supply in the case the unit starts delivering more power than a configured trigger point.

On low-end power supplies based on the half-bridge topology this protection is performed by the PWM controller integrated circuit – as long as it supports it, of course. On power supplies with active PFC circuit, this protection is implemented on the PFC controller.

In both cases what the circuit is really monitoring the total current pulled by the power supply from the power grid. If it increases above a certain value, the

Short-Circuit Protection (SCP)

ELECTRICAL PROTECTIONS Short-circuit protection is probably the oldest form of protection available, being very easy to implement (it is usually implemented outside the monitoring integrated circuit using a couple of transistors).

This is a required protection that will shut down the power supply in the case of any output to “short-circuit,” i.e., to touch the ground line (black wire), either accidentally or in case a component from the computer somehow burns.

No-Load Operation (NLO) Fuse Circuit Breaker PolySwitch RCCB Metal Oxide Varistor Inrush Current Limiter Gas Discharge Tube Spark Gap Lightning ArresterNo-load operation is a required protection that allows the power supply to turn on and work correctly even if there is no load

on its outputs. This is not exactly a “protection” like the ones we’ve seen so far, but more like a design requirement.

1. Open, short, and simultaneous circuits 2. Current Transformers, 1. (CT) Construction and Design, 2. CT Models, 3. CT Saturation, 4. CT Circuit Testing, 5. CT Insulation, polarity, ratio, excitation, winding and lead resistance and burden measurement, testing.

3. ensors – Current Transformers (CT) and Voltage Transformer 4. Protective Relaying Devices 5. . Transformer Differential Relay Protection 6. Breaker Failure Relay Protection 7. Industrial and Commercial Protection 8. Power Component Thermal Protection 9. Digital Relay Protection 10.

The following outline describes the topic material that can be presented to electrical engineers to introduce them to sys-tem protection. It is a comprehensive list of skills needed for an engineer charged with the responsibility for power system protection and control including associated automation func-tions.

B.1 Program content I. Phasor Notation and Network Modeling II. Symmetrical Components III. Sensors – Current Transformers (CT) and Voltage Transformer Theory IV. Protective Relaying Devices V. Settings and Coordination of High Voltage Trans-mission and Subtransmission System Back-up Ground Relaying VI. Setting and Coordination of High Voltage Transmis-sion and Subtransmission Step Distance Phase Re-lays VII. Transformer Differential Relay Protection VIII. Protection for a Transmission/Distribution Substation IX. Breaker Failure Relay Protection X. High Speed Relaying for High Voltage Systems XI. High Speed Relaying for Extra High Voltage Sys-tems XII. Generating Station Equipment Protection XIII. Industrial and Commercial Protection XIV. Power Component Thermal Protection

XV. Power System Stability and Relaying XVI. Advances in Substation and Line Protection XVII. Wide Area System Protection XVIII. Relay System Testing Practical experience may be gained by performing several laboratory experiments. The topics could include, but not limited to, the following topics: A. Short Circuit and Symmetrical Component Fault Calcu-lation Program B. Power System Protection Design and Setting C. Relay Testing D. Substation Control System Design E. Generator and Motor Protection F. IED‟s Configuration and Setting G. Relaying for Stability H. Open Loop and Closed Loop testing using Real-Time Digital Simulation Technology B.2 Detailed Outline I. Phasor Notation and Network Modeling A. Voltage and Current Conventions in a Three Phase System 1. Phasors and the Operators “j” and “a” 2. Percent and Per Unit Calculations

B. Three Phase Transformer Modeling 1. Transformer Delta and Wye Connections 2. Conversion of Delta Impedances to Equivalent Wye Impedances for Three Winding Trans-formers 3. Transformer Impedance Calculations from Nameplate Data C. Transmission Line Impedance Modeling D. Delta Network to Wye Network Conversions II. Symmetrical Components A. Symmetrical Component Equations and Network Models 1. Convert Unbalance Phasors into Symmetrical Components 2. Development of Sequence Networks 3. Sequence Current Flow Through a 3-Phase Transformer including relationship to Phase Shifts 4. Generator Impedance Models 5. Transmission Line Models 6. Sequence Network Modeling Example B. Symmetrical Component Analysis of an Unbalanced System 1. Symmetrical Component Network Connections by Type of Fault 2. Arc Resistance modeling 3. Radial System Fault Current Example 4. Network System Fault Current Examples

5. Open, short, and simultaneous circuits D. Symmetrical Component Computer Programs 1. Short circuit calculation programs 2. Relay coordination computer programs 3. Relay coordination curve plotting programs E. Symmetrical Components as Measured by Relays 1. 3I0 and 3V0 Filters 2. Negative Sequence Voltage and Current Filters 3. Mixed Excitation (I2 –I1/K) Current Filter for Phase Comparison 4. Other Examples of Composite Sequence Filters III. Sensors - Current Transformer and Voltage Transformer Theory A. Current Transformers 1. (CT) Construction and Design 2. CT Models 3. CT Saturation 4. CT Circuit Testing 5. CT Insulation, polarity, ratio, excitation, wind-ing and lead resistance and burden measurement testing B. Capacitive Voltage Transformer (CVT) 1. Construction and Design

2. CVT Models 3. Transient Responses of a CVT 4. CVT Testing C. Non-conventional/Optical instrument transformers 1. Performance and conformance IV. Protective Relaying Devices A. Device Function Numbers and the Need for Stand-ardizing Protection Systems B. Stand Alone Relay Principles of Operation C. Polarized and Direction Ground Overcurrent Re-lays D. Polyphase Distance Protection E. Distance Ground Electromechanical F. Methods of Phase and Ground Impedance Relay Polarization G. Static Phase and Ground Impedance H. High Impedance Bus Differential I. Transformer Differential J. Breaker Failure Relays K. Digital Relay Protection 1. Digital Integrated Circuit Design 2. Digital Systems Design 3. Digital Signal Processing and Filtering

4. Techniques 5. Limitations of Protection 6. Settings Examples 7. Firmware Revision Control L. Transmission Line Teleprotection Communication Media 1. Lease Phone Lines 2. Direct of Fiber Optics 3. Sonet Based Multiplexing 4. Power Line Carrier V. Settings and Coordination of High Voltage Transmission and Subtransmission System Back-up Ground Relaying A. Current and Voltage Polarizing Circuits and Sources B. Effect of Mutual Induction on Ground Relaying C. 138 kV Back-up Ground Relay Settings Example D. Impacts of mis-coordination VI. Settings and Coordination of High Voltage Transmis-sion and Subtransmission Step Distance Phase Relaying A. Introduction to Phase Relay Principles of Operation B. General Torque Equations

C. R-X Diagrams and Phase Impedance Relays D. Relay Loadability/Load Encroachment E. Phase Relay Settings Theory F. Theory of Apparent Impedance Calculations G. Step Distance Phase Relay Settings Example H. Impact of series compensation in impedance computations, protection set points, and coordination I. Special Coordination Considerations 1. Two adjacent line sections with one being very long and the other being very short VII. Transformer Differential Relay Protection A. Two and Three Winding 3 Phase Transformers B. Phase Angle Regulating Transformers (PARS) VIII. Protection for a Transmission/Distribution Substa-tion A. Protection of Transformers 1. Basics of Transformer Design 2. Differential Relay Protection 3. Sudden Pressure Relay 4. Transformer Loading Philosophy 5. Transformer and Bus Back-up Relaying 6. Transformer Automatic Paralleling Circuitry

B. 12.5 kV radial feeder protection 1. Overcurrent relay settings 2. Fuse coordination C. Interlocking Scheme Logic to Facilitate Trans-former Loading and Equipment Restoration D. Breaker Failure Protection E. Circuit Switcher Protection F. Undervoltage and Underfrequency Load shedding Applications and Schemes G. Substation Equipment Thermal and Overload Pro-tection H. Capacitor Bank Protection IX. Breaker Failure Relay Protections A. Circuit Breaker Mechanisms B. Breaker Failure Relay Circuit Designs C. Breaker Failure Relay Timer Setting Considerations D. Variations in Breaker Failure Relaying for HV and EHV Applications X. High Speed Relaying for High Voltage Systems A. High Speed Relaying Philosophies for High Voltage Systems B. Directional Comparison Relay Fundamentals C. Direct Transfer Trip

D. Permissive Overreach Transfer Trip E. Current Differential Transfer Trip F. Traveling wave theory applied to line protection G. Automatic Reclosing H. Protection Requirements in an Urban 138 kV Cable Network 1. Limitation of fault damage 2. Need for backup relay coordination I. Local versus Remote Breaker Failure Protection J. Relay Loadability and Dependability XI. EHV Transmission System Protection A. Protection Philosophy B. Two Systems of Relaying 1. Redundancy advantages and disadvantages 2. Current Only High Speed Relaying 3. Phase and Ground Distance Relaying 4. Impedance Based Direct Underreaching Line Transfer Trip C. High Speed Communication Systems 1. Power Line Carrier 2. Microwave 3. Fiber Optics

4. Relay Loadability D. Fully Complimentary High Speed Scheme Logic E. Stability Ramifications of Relaying F. EHV Line Characteristics G. Effect of line capacitance on relaying H. Autotransformer Protection 1. Differential Protection 2. Polarizing Current Sources 3. Transformer and 138 kV System Backup pro-tection 4. Transformer Relay loadability 5. Transformer Transfer Trip XII. Generating Station Equipment Protection A. Configurations of Generating Stations B. Auxiliary Power Protection 1. Fault Current Calculations at Medium and Low Voltage Levels 2. Low Voltage Protective Devices 3. Low Voltage Unit Substation and Motor Con-trol Center Protection 4. Medium Voltage Protection for Auxiliary Pow-er Unit Substations 5. Medium Voltage Motor Protection a. Motor Design

b. Induction Motors c. Principles of Synchronous Machines d. Variable Speed Motors e. Speed vs Torque Curves f. Thermal Limit Curves g. Locked Rotor h. Duty Factor Ratings of Motors i. Current and Impedance Relay Applications j. Motor Protection Settings k. Specialty Application Motors 6. Medium Voltage Bus Protection 7. Diesel Generator Protection 8. Auxiliary Power System Settings and Coordination Examples 9. Settings and Coordination Computer Programs 10. Nuclear Station Safety Related Protection Requirements C. Generating Unit Protection 1. IEEE Standard C37.102 and the need for Standard Unit Protection 2. Device Numbers and Function 3. Need for system coordination

a. Impacts of miscoordination b. Impacts of overexcitation protection during c. system swings D. Protection Guide for Units and Auxiliaries a. High Voltage connected units b. EHV Connected units c. Independent Power Producers XIII. Unit Transformer Protection XIV. Generator Protective Functions A. Generator Differential B. Stator Ground Fault A. 3V0 Methods B. Neutral Transformer Protection Methods for Stator Protection C. 100% Stator Ground Fault Protection Methods 3. Overexcitation Protection 4. Negative Sequence Protection 5. Exciter Circuitry Basics and Protection 6. Loss of Field Protection 7. Reverse Power Turbine Protection

8. Load Rejection 9. Inadvertent Energization 10. Underfrequency 11. Out-of-Step Protection 12. Overall Settings Examples 13. Integrated Digital Protection Systems 14. Generating Station Interface Protection XV.Power System Stability and Relaying A. Power Angle Equation 1. Simplified Power Angle Curve 2. Equal Area Criteria B. Solution of Power Swing Equation and Inertia 1. General Equation and piecewise linear solution 2. Stable swing studies (units, in particular) C. Steady State and Dynamic Stability D. Transient Stability under Balance Fault Condi-tions E. Multi-Machine Stability F. Underfrequency Load Shedding Protection G. Effects of Protection Systems of Stability

H. Breaker Failure Protection as related to Stability I. Relay Schemes to Maintain System Stability J. Critical Clearing Time Determination K. Unit Stability Tripping 1. Bus Sectionalizing as Related 2. to Stability 3. Load Rejection Tripping 4. Multiple Line Outage Unit 5. Trip Schemes 6. Fault and Path Mode Schemes A. Impact of Automatic Reclosing on System Stability 1. Synchronism Check 2. High Speed Reclosing 3. System Swing Detection 4. Impedance based detection XVI. Industrial and Commercial Protection A. Interface Protection B. Transformer Protection C. Distributed Generation Protection

D. Equipment Protection XVII. Wide Area System Protection A. Remedial Action Schemes B. Power System Stressed Conditions and associated challenges Stability (thermal or angular), voltage collapse, etc.

C. Wide Areas Measurement Systems D. Phasor Displays and Phasor Measurements E. Wide Area Control and Advance Warning Systems F. Total Visibility G. Possible considerations for automatic recovery sys-tems when feasible - Application needs to be cor-rect H. Prevention of System Cascades and Blackouts I. Real Time Predictors of Cascades – Phase Angle and Voltage J. System Restoration Protection Issues XVIII.Power components thermal protection A. Thermal modules for Hot spot monitoring of power transformers; B. Microprocessor based relay for overhead lines thermal protection; C. Power cables overload protection units based on Distributed Temperature Sensors XIX.Advances in Substation and Line Protection A. Substation Automation Communication Meth-ods

1. Automation fundamentals and understanding of basics of interface protocol for use in protection and associate interlocking controls.

2. What is ModBus, DNP, and UCA/IEC-61850 3. Inter-substation communication methods for da-ta gathering and integration 4. Inter-substation communication methods for tripping and closing of circuit breakers 5. Network communication and traffic manage-ment - Industry trend for communication based networking (either at the substation or between substations) 6. IED‟s managed by wireless communication links (i.e. employing GSM/GPRS or satellite based communication units).

B. Substation LAN Management C. Impacts of GPS input selection – Understanding of modulated vs. de-modulated signals.

XX.Relay System Testing A. Stand Alone Relay Tests using Phase Shifter Test Kit B. Substation and Transmission Line Protection Func-tional Testing C. Automatic testing and troubleshooting D. System Testing and hidden failures (manufacturing, relay settings, coordination studies, etc.)

XXI.Protection Lab

A. Short Circuit and Symmetrical Component Fault Calculation Program B. Requires a short circuit computer program, a relay coordination and curve drawing program and a cad program C. Power System Protection Design and Settings Mod-ule D. Requires a short circuit computer program, a relay coordination and curve drawing program and a cad program E. Relay Testing Module F. Requires electromechanical phase and overcurrent relays, Digital Relay (transmission line preferred) and a three phase test instrument G. Substation Control System Design Module H. Requires an digital relay and programmable logic controller I. Generator and Motor Protection Module J. Requires a digital relay, three phase current and voltage test device and rotating machine simulator K. IED Configuration and Settings Module L. Requires a digital relay and three phase test instru-ment M. Relaying For Stability Module N. Requires a system simulator O. Open loop and closed looped testing using Real-time Digital simulation technology

ELECTRÓNICA DE POTENCIA/ ELECTRÓNICA INDUSTRIAL 1 Y LABORATORIO, 2547820/2507385 Trabajo grupal 1,10%, carga r, control por fase, disparo aislado de scr,, entrada de consignas, tiempo ida, tiempo regreso por teclado de pc. septiembre 22 inicio.

1.

Trabajo grupal 2,10%, accionamiento completo, motor BALDOR, DC, excitación independiente, técnica PWM, ingreso de consignas desde plataforma computacional. Entrega

2.

Trabajo grupal 3,10%. Funcionamiento, topología, cálculo,, utilidad fr protecciones: OVP para voltaje de salida variable del tipo CROWBAR , OCP tipo FEEDFORWARD en fuente DC, OCP del tipo FEEDBACK o FOLDBACK en fuente DC, OCP del

tipo SHUTDOWNP Con sensado diferencial de corriente de entrada. Entregar 1 de diciembre.(8días) 3.

Eamen grupal #1 /2020-2/15% 4. Eamen grupal #2 /2020-2/15%.

5.

Simulación grupal, SPICE circuito, 15%

6.

Simulación grupal, LABVIEW, formas de onda 15% 7. Seguimiento y/o emulación de laboratorio, 10%

“Feed-Forward and Feed-Back Circuits of the NRF2/AP-1 Composite Pathway

By Peter V. Zolotukhin and Anna A. Belanova

Submitted: April 19th 2016Reviewed: August 11th 2016Published: December 21st 2016

DOI: 10.5772/65216

Home > Books > A Master Regulator of Oxidative Stress - The Transcription Factor Nrf2

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Abstract

Being the central regulator of oxidative status of the cell, NRF2 must be regulated so that its activity can be rapidly and strongly induced when needed and quickly suppressed when not. Moreover, for the cell, NRF2 means much more than just antioxidant defense.

Numerous general functions rely on NRF2 and related factors. All this implies that the NRF2 pathway has peculiar and powerful mechanisms of control of its activity. To a great extent, these mechanisms are based on feed-forward and feed-back circuits.

These circuits, more than a dozen, are in the focus of this chapter.

“Keywords feed-back regulationfeed-forward regulationSQSTM1SESN2thioredoxinNF-kappaBGSTP1NAPDH oxidaseNRF1NFE2L1NRF3NFE2L3BACH126S proteasomeMIR144mitochondrial biogenesistruncated NRF2NRF2NFE2L2AP-1 Chapter and author infoShow + 1. Introduction

“According to the study by Malhotra et al. [1], Nrf2 controls 1055 protein-coding genes in mice. Although no similar study has ever been performed in humans and given that most routinely studied Nrf2 targets are the same in humans and mice, it is a reasonable assumption that roughly the same number of protein-coding genes is regulated by NRF2 in our species, accounting for astonishing ~5% of all our protein-coding genes. This raises a question: how does NRF2, being a well-known stand-by inducible transcription factor curbed in cytoplasm by KEAP1, fit into cellular context with so many target genes? Among those numerous genes, only a portion codes for immediate antioxidants and detoxifying enzymes. It is the more intriguing since central proteasomal [2], autophagic [3], general signaling [4, 5], and, moreover, cell proliferation, cell cycle and survival regulation factors [1] are in fact NRF2 targets.

Although 645 of the 1055 Nrf2 target genes were found to have basal type of expression regulation by this factor [1], Nrf2 is still required to be present in the nucleus in significant amounts to drive their expression. Notably, most of these genes are not at all related to antioxidant and xenobiotic defense systems, i.e., they are essential for proliferation and basic functioning of the cell. Thus, basal expression driven by NRF2 is important for the cell, but to what extent? One should bear in mind that NRF2 knockouts are nonlethal and generally appear normal in stable laboratory conditions [6], yet they are susceptible to numerous diseases [7–9], especially when the housing conditions are suboptimal [6, 10–12].

This confirms that basal NRF2 functioning, although not vital, is critical for the cell and organism even when no pro-oxidant or xenobiotic exposure is present.

It implies that some cellular mechanisms serve to provide the cell with at least minimal necessary NRF2 activity. On the other hand, it is inherently important for an inducible antioxidant and xenobiotic defense pathway to act as quickly as possible, thus signal amplification is required whenever the initial stimulus is applied.

These two distinct prerequisites for normal cell functioning in everchanging environment appear to be resolved in principle similarly: there are feed-forward circuits that maintain steady-state level of the NRF2 activity and make signal amplification possible when fast cellular reactions are required.

Conversely, too much NRF2 activity is no better than its absence: Keap1-null mice demonstrate postnatal lethality [13, 14], an effect coinciding with nuclear Nrf2 localization [14] and reversed by Nrf2 downregulation [13]. In this case, several feed-back circuits protect the cell and organism from obsessive NRF2 activation.

Most of the feed-forward and feed-back circuits of the NRF2 pathway also involve the closely related transcription factor activator protein 1 (AP-1) represented by homodimers of Jun proteins (JUN, JUNB, JUND) or heterodimers of Fos (FOS, FOSB, FOSL1, FOSL2) and Jun proteins. There are also other dimers of Jun/Fos proteins— e.g. with Atf-proteins [15–17]—still referred to as AP-1 complexes (although either Jun or Fos proteins are absent), and oligomers of Jun/Fos proteins with other proteins. NRF2, in line with NRF1, NRF3 and AP-1 proteins (including Atf proteins) are all basic leucine zipper (bZip) transcription factors of similar structure [18–20].

Functional roles and regulation of Nrf- and AP-1-proteins also significantly overlap [21–24], and, what is even more, these two groups of proteins regulate each other at several levels, including transcription control. It should be especially noted that NRF2 binding site ARE and AP-1 binding site TRE often overlap with AP-1 being embedded into ARE [25, 26]. Having such tight relations, NRF2 and AP-1 obviously form a composite NRF2/AP-1 (or NFE2L2/AP-1) pathway, with both factors contributing much to their own functioning via shared feed-forward and feed-back circuits.

All feed-forward and feed-back circuits are represented, but not outlined or highlighted, in oxidative status interactome map (OSIM) created in our laboratory [26]. This chapter, in contrast, emphasizes on functioning of these circuits and their peculiarities.

2. Feed-forward circuits 2.1. The SESN2/SQSTM1 circuit In 2010, Copple et al., and in 2011, Bui and Shin independently demonstrated that the autophagosomal adaptor protein SQSTM1 (also known as p62) is capable of activating the NRF2 pathway without oxidative modification of KEAP1 protein [27, 28]. This route of the NRF2 pathway activation is one of many known for today.

And by the time of this important discovery, it had already been known that SQSTM1 itself is a target gene of NRF2 [29]. Thus, SQSTM1 was found to form a feed-forward circuit capable of noncanonical activation of NRF2. The mode of action of this factor is in line with its primary function—it merely targets KEAP1 for

autophagosomal degradation, and this process takes place in the cytoplasmic complex of KEAP1 and NRF2 [27, 28]. Copple et al., of immediate relevance to this chapter, also showed that Sqstm1dependent activation of Nrf2 is responsible for ~50% of basal expression of classical Nrf2/Ap-1 targets Nqo1, Gclc and classical Nrf2 target Hmox1 [27].

Later on, Bae et al. [30] revealed that, in mice, Sqstm1 actually promotes Keap1 degradation when Sesn2 is available. A year before, Shin et al. found that Sesn2/SESN2 is also an Nrf2/NRF2 target (Figure 1) [31].

The SESN2/SQSTM1 circuit. Dashed line marks putative interaction.

To our knowledge, to date, there are no direct evidences that the same SESN2/SQSTM1 system is required for KEAP1 degradation.

However, considering evolutionary significance of autophagy and, consequently, SQSTM1, it is highly possible that this holds true for humans as it does for mice. It is also known that SESN2 works together with SQSTM1 in mitophagy, an instance of autophagy [32].

As such, we ourselves have recently tested this hypothesis in our laboratory using the RNA interference approach. We found that in basal conditions SESN2 knockdown caused significant changes in expression of GCLC, HMOX1 and SQSTM1 (all NRF2 targets tested in the study) [33]. Indirect observations emanating from this study also suggest that actually BACH1 expression is also affected,

and BACH1 forms a peculiar feed-back circuit which will be discussed below. In the same study, in pro-oxidant exposure conditions (modeled by culturing HeLa cells in medium containing 400 μM hydrogen peroxide for 24 h after the substance injection; 400 μM hydrogen peroxide was found to be sublethal for HeLa cells in our previous study [34]), on the contrary, SESN2 knockdown caused less pronounced effect on expression of the same set of genes—only HMOX1 and SQSTM1 had significant changes in expression [33]. To the moment of preparation of this chapter, these data are in publication progress.

All together, the existing data clearly show that SESN2/SQSTM1 feed-forward circuit supports basal expression of NRF2/AP-1 targets and to some extent fortifies inducible reaction of this pathway.

2.2. The TXN feed-forward circuit Thioredoxin 1, TXN, has long been known as a critical factor of activation of AP-1 transcription factor [35–37]. This TXN role has an intermediate factor—a DNA repair enzyme and a transcription factor DNA-binding promoter APEX1 [35, 36]. Interestingly, this protein is itself activated by some stimuli that are within primary focus of the NRF2/AP-1 pathway, including pro-oxidants and ionizing radiation [38], but excluding UV radiation [38], as is TXN nuclear import [39, 40].

The TXN feed-forward circuit. Dashed line marks putative interaction.

Years later, Iwasaki et al. found that APEX1 actually facilitates DNA binding to expression of target genes of NRF2 [23] in addition to those of AP-1, NF-kappaB and HIF1A [41]. The same was also confirmed by Shan et al. [42]. As APEX1 reduction by TXN is obligate to its function [37], TXN is a player in NRF2-driven expression, at least in some instances. In this sense, it should be mentioned that the fact that TXN forces NRF2 binding to DNA may be due to direct heterodimerization of Juns (e.g., JUND) with NRF2 [23, 43, 44], as it exactly is in case of FTH1, which was the gene of interest in the study by Iwasaki et al. This raises the possibility that at least in some cases TXN stimulates NRF2 binding to DNA when AP-1 partners are present. Nevertheless, TXN stimulates the NRF2/AP-1 pathway activation.

At the same time, it has already been known for a long time that TXN expression is driven by antioxidant responsive element [45, 46], and Kim et al. directly showed that thioredoxin forces its own expression [22].

TXN functioning as a transcription factor DNA-binding stimulator is conferred by its participation in disulfide-dithiol exchange reactions with APEX1. Once APEX1 has reduced itself in expense of reduced TXN, TXN has to be reduced. There are two major reductases of TXN: TXNRD1 [47] and TXNRD3 [48]. TXNRD1 is in fact an NRF2 target itself [49]. This makes the TXN feed-forward circuit selfsufficient (Figure 2). We found no evidences that TXNRD3 is an

NRF2 target in human, although this is still plausible since this gene/protein is merely poorly studied.

Thioredoxin feed-forward circuit is notable in one more sense: as TRE is often embedded in ARE, one should always consider that an ARE-containing gene may in fact be regulated by AP-1 binding to embedded TRE. This is exactly the case of thioredoxin 1: in their work, Kim et al. found that Jun and Fos overexpression decreased TXN expression in K562 cells [22]. In our work, we have also observed negative interactions between AP-1 and NRF2 working on the same ARE (with embedded TRE) [34], and this mechanism appears to be cell-context dependent, since NRF2 and AP-1 can have both positive and negative effects on the same genes in different cells. This will be discussed in detail in the next section.

Interestingly, TXN feed-forward circuit appears to involve a mechanism of adjustment, which is based on differential control of TXN transcripts. Among NRF2 targets, BACH1 was the first to be described as having individual control of transcripts [50]. Later, we tested whether TXN transcript variants are differentially expressed when the NRF2 pathway gets activated, and so it was [51, 52]. This renders the TXN feed-forward circuit easily adjustable by the cell since only one of the transcripts is NRF2/AP-1 dependent.

The existence of this feed-forward circuit was proven experimentally. However, there are at least two more circuits in the NRF2/AP-1 pathway involving TXN, NRF2 and AP-1. One of them is

a mixed feed-forward/feed-back circuit depending on cellular context (discussed in next section), whereas another is a feed-back circuit (discussed below with other feed-back circuits).

2.3. The AP-1/TXN/NRF2 bidirectional circuit As TXN is an NRF2 target, and TXN is required for AP-1 activation, it forms one more complex circuit. This circuit cannot be considered as either purely feed forward or feed-back. In a set of studies, it was shown that JUN, being a part of AP-1, controls expression of NRF2 [34, 53, 54]. The complexity is that JUN can both activate and suppress the NRF2 expression depending on the cells analyzed: DeNicola et al. [53] demonstrated that, in murine MEF K-RasG12D cells, Jun transactivates Nrf2, whereas Cho et al. [54] and our laboratory [34] showed that JUN has negative influence on NRF2 expression.

Thus, in the cells that are featured by JUN activating NRF2, JUN stimulates expression of NRF2, the latter activates expression of TXN, which, in turn, induces JUN DNA binding. In this case, this circuit acts as feed-forward for both NRF2 and AP-1 (Figure 3A).

In contrast, in cells whose JUN suppresses NRF2, less TXN is expressed and that is less favorable for AP-1 DNA binding, thus releasing suppression from NRF2 expression. Consequently, in this case, the circuit acts as feed-back for AP-1, whereas feed-forward, to some extent, for NRF2 (Figure 3B). It appears that this system favors NRF2 functioning in basal conditions, but acts to suppress it

once NRF2 is activated. We ourselves observed the latter situation [34] but have not yet tested the former.

.

The AP-1/TXN/NRF2 bidirectional circuit. Dashed line marks putative interaction. (A) depicts the situation when cell signaling background determines the system of interaction between NRF2 and AP-1 to act as a feed-forward circuit. On the other hand, (B) illustrates how these interactions form a feed-back circuit in other cell types or cellular conditions.

How is this possible that this circuit differs this much in various cells? The simplest explanation we would suggest is, different cell lines express different amounts of Fos proteins—FOS, FOSB, FOSL1, FOSL2, as well as other Jun proteins—JUNB, JUND [22, 35, 55]. All these proteins significantly differ in their effects on JUN. Moreover, how all Juns perform depends on the presence of JUN dimerizing protein JDP2 capable of deactivating Juns by dimerizing them and by epigenetically silencing the JUN gene [56]. Thus, immediate protein partners of JUN and its powerful regulator JDP2 seem to determine whether JUN will activate or suppress the NRF2 expression in a given cell line. However, this is to be tested in upcoming studies.

2.4. The GSTP1 feed-forward circuit

One of the well-known NRF2 targets, GSTP1 [57, 58], has recently been found to form a novel feed-forward circuit of the NRF2/AP-1 pathway. Carvalho et al. have just recently revealed that, in mice, Gstp1 is capable of S-glutathionylation of Keap1 disrupting the interactions between Keap1 and Nrf2 (Figure 4) [59].

Although this has not been observed in humans to date, it is plausible that the same circuit is also characteristic of the human cell. It is also tempting to suggest that other glutathione-Stransferases may have the same function. At least GSTA1 [60] and GSTA4 [61] expression is, meanwhile, subject to NRF2 regulation.

The GSTP1 feed-forward circuit. GS-S-Keap1—S-glutathionylated Keap1.

Of importance, GSTP1 is activated by both AP-1 and NRF2. Thus, whenever AP-1 is activated, NRF2 is released of KEAP1. This may be a mechanism of protective coupling of cellular events because AP-1 has much more profound effects on the cellular physiology than NRF2 does.

2.5. The NOX4 feed-forward circuit Among peculiar NRF2 targets is NOX4 [62, 63]. This protein is a part of one of NADPH oxidase multiprotein complexes. The inherent NOX4 function is production of superoxide anion, similarly to other Noxes. Two features of this enzyme with respect to other enzymes

of the family are that (1) NOX4 is strongly expressed in kidney where it is suggested to serve as an oxygen sensor for controlling the erythropoietin production [64, 65], and (2) NOX4 is localized to endoplasmic reticulum [66] and nucleus [65] in addition to cell membrane.

Barring NOX5 [67], Noxes require partners for functioning. In case of NOX4, the obligate partner is only one—the CYBA protein (widely known as p22phox) [68]. Thus, the NOX4 feed-forward circuit would require CYBA for functioning, and CYBA is not an NRF2 target.

However, CYBA is in fact an NF-kappaB and AP-1 target [69]. Thus, the composite NRF2/AP-1 pathway controls both enzymes necessary for the NOX4 feed-forward circuit (Figure 5).

The NOX4 feed-forward circuit.

Additionally, NF-kappaB is activated by pro-oxidants [40, 70–72], and DNA binding of both AP-1 and NF-kappaB is stimulated by TXN [40, 70–73].

3. Feed-back circuits As seen from data above, the cellular feed-forward circuits of the NRF2/AP-1 pathway are numerous and versatile providing the cell with required steady-state levels of the proteins, and signal

amplification upon induction of the pathway. However, as already mentioned, hyperactive NRF2 and AP-1 are a death threat to the organism. If NRF2 causes disturbances of tissues functioning, AP-1 is a pro-oncogenic transcription factor (JUN and FOS proteins are also, and in some areas—ordinarily, known as c-jun and c-fos protooncogenes) [74, 75]. In addition to that, NRF2 controls autophagy through SQSTM1, and also transactivates several ABCtransporters, including ABCB1 (MDR1), ABCG2 (BCRP), ABCC2 (MRP2) and ABCC3 (MRP3) [76]—these are all major factors of carcinogenesis and cancer progression. Please note that although autophagy suppresses malignization, once tumor turns malignant, autophagy becomes a cancer-promoting metabolic and drug resistance-conferring trait [77].

Thus, NRF2 and AP-1 activity should be tightly controlled once activated. Several feed-back circuits allow for that. Two of them involve factors participating in feed-forward circuits. These circuits are the first to be discussed.

3.1. The TXN feed-back circuit TXN localizes to cytoplasm and migrates to nucleus upon stimulation. However, the cytoplasmic pool of thioredoxin 1 provides the cell with antioxidant capacity. Thus, cytoplasmic TXN inhibits pro-oxidants-induced NF-kappaB and NRF2 nuclear import [72].

Consequently, as soon as the NRF2/AP-1 pathway activation produces substantial amounts of thioredoxin 1, TXN stabilizes

KEAP1 and promotes NRF2 sequestration and degradation (Figure 6). This works wherever KEAP1 and TXN appear, as will be discussed later.

The TXN feed-back circuit. Dotted line marks an indirect effect.

3.2. The NRF3/NOX4 feed-back circuit One of the least studied, if not the worst studied, bZip proteins is the third homolog of NRF2—NRF3. Very little is known about this transcription factor except the fact that it usually antagonizes NRF1 and NRF2 [22, 78, 79]. Even less is known on stimuli that activate NRF3. NRF3 knockout is known not to change the phenotype to any notable extent [80].

However, Zhang et al. and Pepe et al. revealed that NRF3 is activated by ER stress inducers [81, 82]. Once activated, NRF3, predictably, suppresses ordinary NRF2 target genes [82].

Surprisingly, in contrast to these genes, NOX4 was activated by NRF3 [82]. As NOX4 is an enzyme localized to endoplasmic reticulum and producing superoxide anion, it is capable of oxidative modification of inositol 1,4,5-trisphosphate receptors [83]. Once oxidized, these receptors cause calcium efflux from endoplasmic reticulum to cytosol. This event leads to endoplasmic reticulum stress [83].

As mentioned above, NOX4, for its functioning, requires an AP-1 target—CYBA. ER stress is not a leading, if at all, stimulus for AP-1 activation, yet ER stress causes ROS production (during the protein refolding period), and ROS activation is an acknowledged property of AP-1 [35]. Additionally, NRF3 heterodimerizes with AP-1 proteins FOS and FOSL1 (it was suggested by Zhang et al. that NRF3 thus suppresses TRE-containing NRF2 targets only [81]) and is suggested to heterodimerize with Jun proteins (Figure 7) [79, 84].

The NRF3/NOX4 feed-back circuit. Dashed line marks putative interaction.

The fact that NRF3 activates NOX4 also suggests that the NOX4 feed-forward circuit may equally work for both NRF2 and AP-1.

3.3. The KEAP1 feed-back circuit KEAP1 is the immediate antagonist of NFR2 as it sequesters it and targets it for proteasomal degradation. In a complex experiment, Lee et al. demonstrated that murine Keap1 is another Nrf2 target [85]. Unfortunately, no direct evidences that the same is true for humans are known to the authors to date of preparation of this chapter. This is an extremely important issue to be addressed in future studies. Nevertheless, at least in mice, this circuit is active and facilitates fast Nrf2 pathway shutdown as soon as the oxidative status of the cell is normalized (Figure 8).

Interestingly, KEAP1 can be imported into the nucleus by KPNA6 [46, 60, 86]. In the nucleus, KEAP1 acts exactly as in the cytoplasm—it binds NFE2L2 and induces its polyubiquitination thus targeting it for degradation [46, 85].

It means that GSK3B activity providing NRF2 nuclear export [87–89] is not even required for KEAP1 to force NRF2 degradation.

However, KEAP1 nuclear import is a tightly controlled process. An example of the chain of events causing KEAP1 nuclear import is presented in the next section.

The KEAP1 feed-back circuit.

3.4. The NF-kappaB/KEAP1 feed-back circuit KEAP1 nuclear import stimulation is well described for the case of NF-kappaB activation. Generally, the NF-kappaB and NRF2 pathways antagonize in several ways: NF-kappaB protein RELA (p65) promotes HDAC3 interaction with CREBBP or MAFK thus causing local hypoacetylation surrounding ARE [90]; KEAP1 promotes IKK degradation thus suppressing NF-kappaB release from IKBs [91]; and, in turn, RELA promotes KEAP1 nuclear import [46].

Thus, whenever NF-kappaB is activated, it suppresses NFE2L2 by two mechanisms: by inducing its nuclear sequestration by KEAP1 and by causing transcription-suppressing epigenomic modification

of the NRF2-dependent loci. In this sense, the NF-kappaB/KEAP1 circuit presented here is feed-back for the NRF2/AP-1 pathway yet feed-forward for the NF-kappaB pathway, because less KEAP1 causes less IKK degradation consequently promoting NF-kappaB nuclear import. This feed-forward circuit is probably disrupted by TXN, which is NRF2-dependent (Figure 9).

The NF-kappaB/KEAP1 feed-back circuit.

3.5. The BACH1 feed-back circuit In contrast to the preferentially cytoplasmic NRF2 inhibitor KEAP1, BACH1 is its nuclear antagonist. Just as NRF2, BACH1 belongs to the bZip family, and moreover, to cap‘n’collar (CNC) sub-family.

Thus, BACH1 and NRF2 act in a very similar manner—by heterodimerizing with small Maf proteins [86, 92–94]. There is, however, one striking difference between the two transcription factors: BACH1 acts only on clustered-ARE genes [49, 93], like HMOX1 [93, 95] and NQO1 [96].

As it was discussed earlier, different NRF2 targets may response distinctly to the pathway stimulation or release of pathway suppression. This actually may be caused by BACH1. For example, TXNRD1 is proven to be BACH1-independent [49], while HMOX1 has quadruple ARE [93] being subject to BACH1-induced suppression, and NQO1 has a double ARE still being somewhat suppressed by BACH1 [96]. It appears that BACH1 functioning allows the cell to discriminate between the NRF2 targets. It is

achieved by a relatively simple mechanism: if NRF2 activity is induced by whatever mechanism or stimulus except for pro-oxidant exposure, BACH1 does not allow NRF2 to act on the clustered-ARE genes. In contrast, when the cell is exposed to pro-oxidants of whatever nature, BACH1 is fast oxidized [19, 49, 86, 93, 96, 97], rapidly detaches from DNA [93, 94, 98] and readily degrades [93, 97].

At the same time, BACH1 is itself an NRF2 target and an extremely interesting one. The BACH1 locus codes for three transcript variants produced as the result of alternative transcription initiation. Only BACH1 transcript variant 2 is NRF2-dependent: BACH1 intronic +1411 nt ARE is functional and promotes transcription of the second intron of the gene [50]. The authors of this discovery, Jyrkkänen et al., proposed the existence of the BACH1 feed-back circuit along with their discovery of the functional intronic ARE [50].

The existence of the BACH1 feed-back circuit predisposes different expression dynamics of NRF2 targets depending on their ARE structure: once the NRF2 pathway normalizes oxidative status of the cell, newly synthesized BACH1 protein successfully outcompetes NRF2 on the clustered-ARE loci, while allowing it to act further on non-clustered ARE genes (Figure 10).

The BACH1 feed-back circuit.

In one of our studies, we appeared to observe these events after we released the JUN-imposed suppression of NRF2 in cells treated with hydrogen peroxide: in contrast non-clustered ARE genes (FTH1, CBR3, SQSTM1), HMOX1 expression did not rise— probably because BACH1, being a non-clustered ARE gene, had increased expression in these those settings [34].

3.6. The 26S proteasome/NRF2 feed-back circuit Once polyubiquitinated, NRF2 is degraded by 26S proteasome [86].

Proteasome inhibitors suppress the degradation of NRF2 and stimulate its nuclear import [99].

At the same time, as shown in mice, Nrf2 controls at least eight genes of proteasomal proteins: Psma1, Psma4, Psmb3, Psmb5, Psmb6, Psmc1, Psmc3 and Psmd14 (Figure 11) [2].

Figure 11.

The 26S proteasome/NRF2 feed-back circuit.

To date of preparation of the chapter, no similar results obtained from studies on human cells have been published. Nevertheless, considering the biological significance of the 26S proteasome, it is not an unfair assumption that the same genes are probably controlled by NRF2 in humans.

Notably, even in human cells, AP-1 controls expression of interferon gamma [100], which, in turn, transactivates PSME1 [101] and PSME2 [102] subunit genes of the 26S proteasome.

Thus, NRF2 pathway activation facilitates NRF2 proteasomal degradation once KEAP1 is re-enabled to sequester NRF2. At least, this system works in mice, and probably in humans, too, especially since two Psme genes are under the NRF2/AP-1 pathway control.

3.7. The MIR-144 feed-back circuit One of numerous NRF2-antagonizing miRNAs, MIR-144, is expressed under control of AP-1 [103]. This miRNA not merely blocks translation of NRF2, but rather degrades the NRF2 mRNA [104, 105].

3.8. The mitochondrial bidirectional circuit Being regulated by reactive oxygen species and signaling background of the cell, the NRF2/AP-1 pathway depends on mitochondrial function, including their vast signaling activity [106].

Surprisingly, the NRF2/AP-1 pathway has been found to modulate mitochondrial biogenesis by two mechanisms. The first one has direct experimental evidences: NRF2 is a transcriptional regulator of NRF1—nuclear respiratory factor (which is not to be confused with NFE2L1, an NRF2 homolog, also known and referred to as NRF1 in this chapter) [107]. NRF1, in turn, controls a battery of nuclear genes required for mitochondrial biogenesis and function [107]. The

second mechanism has not been observed in direct experiments and is related to the fact that the NRF1 protein is also controlled by estrogen receptors—a distinct family of transcription factors [108].

However, it is well-known that estrogen receptors act in an intimate collaboration with AP-1 proteins [109, 110]. For example, FOS alone is required for expression of 37% of estrogen receptors target genes [111]. Even though the mitochondrial biogenesis-related targets of the estrogen receptors may not fall into the AP-1-dependent category, which is to be addressed in the future, there is another doubtless interaction between the estrogen receptors and the NRF2/AP-1 pathway. The already discussed NRF2/AP-1 target, TXNRD1 is a strong promoter of transactivatory function of the estrogen receptors [111], thus the NRF2/AP-1 pathway positively regulates estrogen receptors function, while estrogen receptors facilitate mitochondria biogenesis.

Although mitochondrial biogenesis eventually leads to increased ROS generation, this circuit is unlikely to be purely or even mostly feed-forward. Rather, this circuit is bidirectional with strong feedback action because the outer mitochondrial membrane carries the PGAM5 protein. PGAM5, in turn, tethers the NRF2-KEAP1 complexes to mitochondria and acts as a powerful suppressor of the NRF2-dependent expression (Figure 12) [106, 112].

.

The mitochondrial bi-directional circuit. NRF1 here is nuclear respiratory factor 1.

Thus, the more mitochondria, the more KEAP1-NRF2 complexes are likely to be bound to them, leading to decreased availability of NRF2 to nucleus. Unfortunately, it is not known to date how PGAM5 expression is controlled with relation to mitochondria biogenesis— the protein is poorly studied.

It is worth noting that the PGAM5/KEAP1-NRF2 system is suggested to represent a stand-by alarm inducible by changes in mitochondrial function [106, 112]. PGAM5 also appears to stimulate mitophagy as a part of cellular defense against excessive reactive oxygen species [113]. It could be suggested that this function of PGAM5 is directly related to its sequestering of NRF2 with subsequent release of it upon mitochondrial dysfunction: NRF2 controls expression of SQSTM1 and SESN2, two factors of autophagy/mitophagy already discussed above [32, 114].

3.9. The NRF2 bidirectional circuit NRF2 expression is known to respond to stimuli that activate the NRF2/AP-1 pathway [34, 51, 115]. We and others observed that, for reasons still to be revealed, NRF2 had a humped expression curve when different cells had been treated with various pro-oxidant substances [34, 115]. It is possible that this is related to the AP1/TXN/NRF2 bidirectional circuit. It is tempting to suggest that this is caused by AP-1 components that are expressed with different dynamics upon stimulation.

For example, Siriani et al. demonstrated, that angiotensin II, over the same period of treatment, induced 300-fold change in expression of FOS, 500-fold change in expression of FOSB, 2-fold change in expression of FOSL1 and FOSL2, 4-fold change in expression of JUN, 14-fold change in expression of JUNB and 2-fold change in expression of JUND. The same was also fair for the classical AP-1 inducer—TPA: it caused 194-fold change in expression of FOS, 65-fold change in expression of FOSB, 2-fold change in expression of FOSL1, 2-fold change in expression of JUN and JUND, and 9-fold change in expression of JUNB [116].

Since overflow inhibition is characteristic of AP-1 [117], and competition of bZip dimers is a must, it is possible that alterations in ratio of AP-1 components controlling NRF2 expression lead to its humped dosage-expression curve.

This circuit cannot be considered purely feed-back, because before certain threshold, the stimulus activates NRF2 expression. Yet, this character of expression of NRF2 is not entirely of feed-forward nature, and this is discussed in the next, last, section of this chapter.

. The truncated NRF2 feed-back circuit NRF2 is cleaved into a truncated form named truncated NRF2, or tNRF2, by Ice family-like caspases [118]. Just as the full-length

protein, tNRF2 enters nucleus, but there it antagonizes the normal NRF2 and suppresses ARE-containing loci [19, 118].

4. Conclusion Although studying the NRF2/AP-1 pathway targets is an extremely complex task due to their plentitude, experimental evidences suggest that this relatively small, with respect to the regulatory proteins, pathway has enormous number of feed-forward and feedback circuits, some of them being bidirectional.

Some evidences demonstrated directly that these circuits contribute to either fast activation or quenching of the pathway. For other circuits, no such testing has ever been performed. Nevertheless, the existing data unambiguously point out that these circuits are an essential feature of the NRF2/AP-1 pathway. These circuits must be accounted in all applications—starting from cell sensor-based pharmacological screening techniques [119, 120] and ending with therapeutical research and development [121].”

Abstract Being the central regulator of oxidative status of the cell, NRF2 must be regulated so that its activity can be rapidly and strongly induced when needed and quickly suppressed when not. Moreover, for the cell, NRF2 means much more than just antioxidant defense. Numerous general functions rely on NRF2 and related factors. All this implies that the NRF2 pathway has peculiar and powerful mechanisms of control of

its activity. To a great extent, these mechanisms are based on feedforward and feed-back circuits. These circuits, more than a dozen, are in the focus of this chapter.

Chapter and author info Show + 1. Introduction According to the study by Malhotra et al. [1], Nrf2 controls 1055 protein-coding genes in mice. Although no similar study has ever been performed in humans and given that most routinely studied Nrf2 targets are the same in humans and mice, it is a reasonable assumption that roughly the same number of protein-coding genes is regulated by NRF2 in our species, accounting for astonishing ~5% of all our proteincoding genes. This raises a question: how does NRF2, being a wellknown stand-by inducible transcription factor curbed in cytoplasm by KEAP1, fit into cellular context with so many target genes? Among those numerous genes, only a portion codes for immediate antioxidants and detoxifying enzymes. It is the more intriguing since central proteasomal [2], autophagic [3], general signaling [4, 5], and, moreover, cell proliferation, cell cycle and survival regulation factors [1] are in fact NRF2 targets.

Although 645 of the 1055 Nrf2 target genes were found to have basal type of expression regulation by this factor [1], Nrf2 is still required to be present in the nucleus in significant amounts to drive their expression. Notably, most of these genes are not at all related to antioxidant and xenobiotic defense systems, i.e., they are essential for proliferation and basic functioning of the cell. Thus, basal expression driven by NRF2 is important for the cell, but to what extent? One

should bear in mind that NRF2 knockouts are nonlethal and generally appear normal in stable laboratory conditions [6], yet they are susceptible to numerous diseases [7–9], especially when the housing conditions are suboptimal [6, 10–12]. This confirms that basal NRF2 functioning, although not vital, is critical for the cell and organism even when no pro-oxidant or xenobiotic exposure is present.

It implies that some cellular mechanisms serve to provide the cell with at least minimal necessary NRF2 activity. On the other hand, it is inherently important for an inducible antioxidant and xenobiotic defense pathway to act as quickly as possible, thus signal amplification is required whenever the initial stimulus is applied. These two distinct prerequisites for normal cell functioning in ever-changing environment appear to be resolved in principle similarly: there are feed-forward circuits that maintain steady-state level of the NRF2 activity and make signal amplification possible when fast cellular reactions are required.

Conversely, too much NRF2 activity is no better than its absence: Keap1-null mice demonstrate postnatal lethality [13, 14], an effect coinciding with nuclear Nrf2 localization [14] and reversed by Nrf2 downregulation [13]. In this case, several feed-back circuits protect the cell and organism from obsessive NRF2 activation.

Most of the feed-forward and feed-back circuits of the NRF2 pathway also involve the closely related transcription factor activator protein 1 (AP-1) represented by homodimers of Jun proteins (JUN, JUNB, JUND) or heterodimers of Fos (FOS, FOSB, FOSL1, FOSL2) and Jun proteins. There are also other dimers of Jun/Fos proteins—e.g. with Atf-proteins [15–17]—still referred to as AP-1 complexes (although either Jun or Fos proteins are absent), and oligomers of Jun/Fos proteins with other proteins. NRF2, in line with NRF1, NRF3 and AP-1

proteins (including Atf proteins) are all basic leucine zipper (bZip) transcription factors of similar structure [18–20]. Functional roles and regulation of Nrf- and AP-1-proteins also significantly overlap [21–24], and, what is even more, these two groups of proteins regulate each other at several levels, including transcription control. It should be especially noted that NRF2 binding site ARE and AP-1 binding site TRE often overlap with AP-1 being embedded into ARE [25, 26].

Having such tight relations, NRF2 and AP-1 obviously form a composite NRF2/AP-1 (or NFE2L2/AP-1) pathway, with both factors contributing much to their own functioning via shared feed-forward and feed-back circuits.

All feed-forward and feed-back circuits are represented, but not outlined or highlighted, in oxidative status interactome map (OSIM) created in our laboratory [26]. This chapter, in contrast, emphasizes on functioning of these circuits and their peculiarities.

2. Feed-forward circuits 2.1. The SESN2/SQSTM1 circuit In 2010, Copple et al., and in 2011, Bui and Shin independently demonstrated that the autophagosomal adaptor protein SQSTM1 (also known as p62) is capable of activating the NRF2 pathway without oxidative modification of KEAP1 protein [27, 28]. This route of the NRF2 pathway activation is one of many known for today. And by the time of this important discovery, it had already been known that SQSTM1 itself is a target gene of NRF2 [29]. Thus, SQSTM1 was found to form a feed-forward circuit capable of noncanonical activation of NRF2. The mode of action of this factor is in line with its primary function—it merely targets KEAP1 for autophagosomal degradation,

and this process takes place in the cytoplasmic complex of KEAP1 and NRF2 [27, 28]. Copple et al., of immediate relevance to this chapter, also showed that Sqstm1-dependent activation of Nrf2 is responsible for ~50% of basal expression of classical Nrf2/Ap-1 targets Nqo1, Gclc and classical Nrf2 target Hmox1 [27].

Later on, Bae et al. [30] revealed that, in mice, Sqstm1 actually promotes Keap1 degradation when Sesn2 is available. A year before, Shin et al. found that Sesn2/SESN2 is also an Nrf2/NRF2 target (Figure 1) [31].

Figure 1.

The SESN2/SQSTM1 circuit. Dashed line marks putative interaction.

To our knowledge, to date, there are no direct evidences that the same SESN2/SQSTM1 system is required for KEAP1 degradation. However, considering evolutionary significance of autophagy and, consequently, SQSTM1, it is highly possible that this holds true for humans as it does for mice. It is also known that SESN2 works together with SQSTM1 in

mitophagy, an instance of autophagy [32]. As such, we ourselves have recently tested this hypothesis in our laboratory using the RNA interference approach. We found that in basal conditions SESN2 knockdown caused significant changes in expression of GCLC, HMOX1 and SQSTM1 (all NRF2 targets tested in the study) [33]. Indirect observations emanating from this study also suggest that actually BACH1 expression is also affected, and BACH1 forms a peculiar feed-back circuit which will be discussed below. In the same study, in pro-oxidant exposure conditions (modeled by culturing HeLa cells in medium containing 400 μM hydrogen peroxide for 24 h after the substance injection; 400 μM hydrogen peroxide was found to be sublethal for HeLa cells in our previous study [34]), on the contrary, SESN2 knockdown caused less pronounced effect on expression of the same set of genes—only HMOX1 and SQSTM1 had significant changes in expression [33]. To the moment of preparation of this chapter, these data are in publication progress.

All together, the existing data clearly show that SESN2/SQSTM1 feedforward circuit supports basal expression of NRF2/AP-1 targets and to some extent fortifies inducible reaction of this pathway.

2.2. The TXN feed-forward circuit Thioredoxin 1, TXN, has long been known as a critical factor of activation of AP-1 transcription factor [35–37]. This TXN role has an intermediate factor—a DNA repair enzyme and a transcription factor DNA-binding promoter APEX1 [35, 36]. Interestingly, this protein is itself activated by some stimuli that are within primary focus of the NRF2/AP-1 pathway, including pro-oxidants and ionizing radiation [38], but excluding UV radiation [38], as is TXN nuclear import [39, 40].

Figure 2.

The TXN feed-forward circuit. Dashed line marks putative interaction.

Years later, Iwasaki et al. found that APEX1 actually facilitates DNA binding to expression of target genes of NRF2 [23] in addition to those of AP-1, NF-kappaB and HIF1A [41]. The same was also confirmed by Shan et al. [42]. As APEX1 reduction by TXN is obligate to its function [37], TXN is a player in NRF2-driven expression, at least in some instances. In this sense, it should be mentioned that the fact that TXN forces NRF2 binding to DNA may be due to direct heterodimerization of Juns (e.g., JUND) with NRF2 [23, 43, 44], as it exactly is in case of FTH1, which was the gene of interest in the study by Iwasaki et al.

This raises the possibility that at least in some cases TXN stimulates NRF2 binding to DNA when AP-1 partners are present. Nevertheless, TXN stimulates the NRF2/AP-1 pathway activation.

At the same time, it has already been known for a long time that TXN expression is driven by antioxidant responsive element [45, 46], and Kim et al. directly showed that thioredoxin forces its own expression [22].

TXN functioning as a transcription factor DNA-binding stimulator is conferred by its participation in disulfide-dithiol exchange reactions with APEX1. Once APEX1 has reduced itself in expense of reduced TXN, TXN has to be reduced. There are two major reductases of TXN: TXNRD1 [47] and TXNRD3 [48]. TXNRD1 is in fact an NRF2 target itself [49]. This makes the TXN feed-forward circuit self-sufficient (Figure 2). We found no evidences that TXNRD3 is an NRF2 target in human, although this is still plausible since this gene/protein is merely poorly studied.

Thioredoxin feed-forward circuit is notable in one more sense: as TRE is often embedded in ARE, one should always consider that an AREcontaining gene may in fact be regulated by AP-1 binding to embedded TRE. This is exactly the case of thioredoxin 1: in their work, Kim et al. found that Jun and Fos overexpression decreased TXN expression in K562 cells [22]. In our work, we have also observed negative interactions between AP-1 and NRF2 working on the same ARE (with embedded TRE) [34], and this mechanism appears to be cell-context dependent, since NRF2 and AP-1 can have both positive and negative effects on the same genes in different cells. This will be discussed in detail in the next section.

Interestingly, TXN feed-forward circuit appears to involve a mechanism of adjustment, which is based on differential control of TXN transcripts.

Among NRF2 targets, BACH1 was the first to be described as having individual control of transcripts [50]. Later, we tested whether TXN transcript variants are differentially expressed when the NRF2 pathway gets activated, and so it was [51, 52]. This renders the TXN feed-forward circuit easily adjustable by the cell since only one of the transcripts is NRF2/AP-1 dependent.

The existence of this feed-forward circuit was proven experimentally.

However, there are at least two more circuits in the NRF2/AP-1 pathway involving TXN, NRF2 and AP-1. One of them is a mixed feedforward/feed-back circuit depending on cellular context (discussed in next section), whereas another is a feed-back circuit (discussed below with other feed-back circuits).

2.3. The AP-1/TXN/NRF2 bidirectional circuit As TXN is an NRF2 target, and TXN is required for AP-1 activation, it forms one more complex circuit. This circuit cannot be considered as either purely feed forward or feed-back. In a set of studies, it was shown that JUN, being a part of AP-1, controls expression of NRF2 [34, 53, 54]. The complexity is that JUN can both activate and suppress the NRF2 expression depending on the cells analyzed: DeNicola et al. [53] demonstrated that, in murine MEF K-RasG12D cells, Jun transactivates Nrf2, whereas Cho et al. [54] and our laboratory [34] showed that JUN has negative influence on NRF2 expression.

Thus, in the cells that are featured by JUN activating NRF2, JUN stimulates expression of NRF2, the latter activates expression of TXN,

which, in turn, induces JUN DNA binding. In this case, this circuit acts as feed-forward for both NRF2 and AP-1 (Figure 3A).

In contrast, in cells whose JUN suppresses NRF2, less TXN is expressed and that is less favorable for AP-1 DNA binding, thus releasing suppression from NRF2 expression. Consequently, in this case, the circuit acts as feed-back for AP-1, whereas feed-forward, to some extent, for NRF2 (Figure 3B). It appears that this system favors NRF2 functioning in basal conditions, but acts to suppress it once NRF2 is activated. We ourselves observed the latter situation [34] but have not yet tested the former.

Figure 3.

The AP-1/TXN/NRF2 bidirectional circuit. Dashed line marks putative interaction. (A) depicts the situation when cell signaling background determines the system of interaction between NRF2 and AP-1 to act as a feed-forward circuit. On the other hand, (B) illustrates how these interactions form a feed-back circuit in other cell types or cellular conditions.

How is this possible that this circuit differs this much in various cells?

The simplest explanation we would suggest is, different cell lines express different amounts of Fos proteins—FOS, FOSB, FOSL1, FOSL2, as well as other Jun proteins—JUNB, JUND [22, 35, 55]. All these proteins significantly differ in their effects on JUN. Moreover, how all Juns perform depends on the presence of JUN dimerizing protein JDP2 capable of deactivating Juns by dimerizing them and by epigenetically silencing the JUN gene [56]. Thus, immediate protein partners of JUN and its powerful regulator JDP2 seem to determine whether JUN will activate or suppress the NRF2 expression in a given cell line. However, this is to be tested in upcoming studies.

2.4. The GSTP1 feed-forward circuit One of the well-known NRF2 targets, GSTP1 [57, 58], has recently been found to form a novel feed-forward circuit of the NRF2/AP-1 pathway. Carvalho et al. have just recently revealed that, in mice, Gstp1 is capable of S-glutathionylation of Keap1 disrupting the interactions between Keap1 and Nrf2 (Figure 4) [59].

Although this has not been observed in humans to date, it is plausible that the same circuit is also characteristic of the human cell. It is also tempting to suggest that other glutathione-S-transferases may have the same function. At least GSTA1 [60] and GSTA4 [61] expression is, meanwhile, subject to NRF2 regulation.

Figure 4.

The GSTP1 feed-forward circuit. GS-S-Keap1—S-glutathionylated Keap1.

Of importance, GSTP1 is activated by both AP-1 and NRF2. Thus, whenever AP-1 is activated, NRF2 is released of KEAP1. This may be a mechanism of protective coupling of cellular events because AP-1 has much more profound effects on the cellular physiology than NRF2 does.

2.5. The NOX4 feed-forward circuit Among peculiar NRF2 targets is NOX4 [62, 63]. This protein is a part of one of NADPH oxidase multiprotein complexes. The inherent NOX4 function is production of superoxide anion, similarly to other Noxes.

Two features of this enzyme with respect to other enzymes of the family are that (1) NOX4 is strongly expressed in kidney where it is suggested to serve as an oxygen sensor for controlling the

erythropoietin production [64, 65], and (2) NOX4 is localized to endoplasmic reticulum [66] and nucleus [65] in addition to cell membrane.

Barring NOX5 [67], Noxes require partners for functioning. In case of NOX4, the obligate partner is only one—the CYBA protein (widely known as p22phox) [68]. Thus, the NOX4 feed-forward circuit would require CYBA for functioning, and CYBA is not an NRF2 target.

However, CYBA is in fact an NF-kappaB and AP-1 target [69]. Thus, the composite NRF2/AP-1 pathway controls both enzymes necessary for the NOX4 feed-forward circuit (Figure 5).

Figure 5.

The NOX4 feed-forward circuit.

Additionally, NF-kappaB is activated by pro-oxidants [40, 70–72], and DNA binding of both AP-1 and NF-kappaB is stimulated by TXN [40, 70–73].

3. Feed-back circuits As seen from data above, the cellular feed-forward circuits of the NRF2/AP-1 pathway are numerous and versatile providing the cell with required steady-state levels of the proteins, and signal amplification upon induction of the pathway. However, as already mentioned, hyperactive NRF2 and AP-1 are a death threat to the organism. If NRF2 causes disturbances of tissues functioning, AP-1 is a prooncogenic transcription factor (JUN and FOS proteins are also, and in some areas—ordinarily, known as c-jun and c-fos proto-oncogenes) [74, 75]. In addition to that, NRF2 controls autophagy through SQSTM1, and also transactivates several ABC-transporters, including ABCB1 (MDR1), ABCG2 (BCRP), ABCC2 (MRP2) and ABCC3 (MRP3) [76]—these are all major factors of carcinogenesis and cancer progression. Please note that although autophagy suppresses malignization, once tumor turns malignant, autophagy becomes a cancer-promoting metabolic and drug resistance-conferring trait [77].

Thus, NRF2 and AP-1 activity should be tightly controlled once activated. Several feed-back circuits allow for that. Two of them involve factors participating in feed-forward circuits. These circuits are the first to be discussed.

3.1. The TXN feed-back circuit TXN localizes to cytoplasm and migrates to nucleus upon stimulation.

However, the cytoplasmic pool of thioredoxin 1 provides the cell with antioxidant capacity. Thus, cytoplasmic TXN inhibits pro-oxidantsinduced NF-kappaB and NRF2 nuclear import [72].

Consequently, as soon as the NRF2/AP-1 pathway activation produces substantial amounts of thioredoxin 1, TXN stabilizes KEAP1 and promotes NRF2 sequestration and degradation (Figure 6). This works wherever KEAP1 and TXN appear, as will be discussed later.

Figure 6.

The TXN feed-back circuit. Dotted line marks an indirect effect.

3.2. The NRF3/NOX4 feed-back circuit One of the least studied, if not the worst studied, bZip proteins is the third homolog of NRF2—NRF3. Very little is known about this transcription factor except the fact that it usually antagonizes NRF1 and NRF2 [22, 78, 79]. Even less is known on stimuli that activate NRF3. NRF3 knockout is known not to change the phenotype to any notable extent [80].

However, Zhang et al. and Pepe et al. revealed that NRF3 is activated

by ER stress inducers [81, 82]. Once activated, NRF3, predictably,

suppresses ordinary NRF2 target genes [82]. Surprisingly, in contrast to these genes, NOX4 was activated by NRF3 [82]. As NOX4 is an enzyme localized to endoplasmic reticulum and producing superoxide anion, it is capable of oxidative modification of inositol 1,4,5trisphosphate receptors [83]. Once oxidized, these receptors cause calcium efflux from endoplasmic reticulum to cytosol. This event leads to endoplasmic reticulum stress [83].

As mentioned above, NOX4, for its functioning, requires an AP-1 target—CYBA. ER stress is not a leading, if at all, stimulus for AP-1 activation, yet ER stress causes ROS production (during the protein refolding period), and ROS activation is an acknowledged property of AP-1 [35]. Additionally, NRF3 heterodimerizes with AP-1 proteins FOS and FOSL1 (it was suggested by Zhang et al. that NRF3 thus suppresses TRE-containing NRF2 targets only [81]) and is suggested to heterodimerize with Jun proteins (Figure 7) [79, 84].

Figure 7.

The NRF3/NOX4 feed-back circuit. Dashed line marks putative interaction.

The fact that NRF3 activates NOX4 also suggests that the NOX4 feedforward circuit may equally work for both NRF2 and AP-1.

3.3. The KEAP1 feed-back circuit KEAP1 is the immediate antagonist of NFR2 as it sequesters it and targets it for proteasomal degradation. In a complex experiment, Lee et al. demonstrated that murine Keap1 is another Nrf2 target [85].

Unfortunately, no direct evidences that the same is true for humans are known to the authors to date of preparation of this chapter. This is an extremely important issue to be addressed in future studies.

Nevertheless, at least in mice, this circuit is active and facilitates fast Nrf2 pathway shutdown as soon as the oxidative status of the cell is normalized (Figure 8).

Interestingly, KEAP1 can be imported into the nucleus by KPNA6 [46, 60, 86]. In the nucleus, KEAP1 acts exactly as in the cytoplasm—it binds NFE2L2 and induces its polyubiquitination thus targeting it for degradation [46, 85].

It means that GSK3B activity providing NRF2 nuclear export [87–89] is not even required for KEAP1 to force NRF2 degradation. However, KEAP1 nuclear import is a tightly controlled process. An example of the chain of events causing KEAP1 nuclear import is presented in the next section.

Figure 8.

The KEAP1 feed-back circuit.

3.4. The NF-kappaB/KEAP1 feed-back circuit KEAP1 nuclear import stimulation is well described for the case of NFkappaB activation. Generally, the NF-kappaB and NRF2 pathways antagonize in several ways: NF-kappaB protein RELA (p65) promotes HDAC3 interaction with CREBBP or MAFK thus causing local hypoacetylation surrounding ARE [90]; KEAP1 promotes IKK degradation thus suppressing NF-kappaB release from IKBs [91]; and, in turn, RELA promotes KEAP1 nuclear import [46].

Thus, whenever NF-kappaB is activated, it suppresses NFE2L2 by two mechanisms: by inducing its nuclear sequestration by KEAP1 and by causing transcription-suppressing epigenomic modification of the NRF2-dependent loci. In this sense, the NF-kappaB/KEAP1 circuit presented here is feed-back for the NRF2/AP-1 pathway yet feedforward for the NF-kappaB pathway, because less KEAP1 causes less IKK degradation consequently promoting NF-kappaB nuclear import.

This feed-forward circuit is probably disrupted by TXN, which is NRF2dependent (Figure 9).

Figure 9.

The NF-kappaB/KEAP1 feed-back circuit.

3.5. The BACH1 feed-back circuit In contrast to the preferentially cytoplasmic NRF2 inhibitor KEAP1, BACH1 is its nuclear antagonist. Just as NRF2, BACH1 belongs to the bZip family, and moreover, to cap‘n’collar (CNC) sub-family. Thus, BACH1 and NRF2 act in a very similar manner—by heterodimerizing with small Maf proteins [86, 92–94]. There is, however, one striking difference between the two transcription factors: BACH1 acts only on clustered-ARE genes [49, 93], like HMOX1 [93, 95] and NQO1 [96].

As it was discussed earlier, different NRF2 targets may response distinctly to the pathway stimulation or release of pathway suppression.

This actually may be caused by BACH1. For example, TXNRD1 is

proven to be BACH1-independent [49], while HMOX1 has quadruple ARE [93] being subject to BACH1-induced suppression, and NQO1 has a double ARE still being somewhat suppressed by BACH1 [96]. It appears that BACH1 functioning allows the cell to discriminate between the NRF2 targets. It is achieved by a relatively simple mechanism: if NRF2 activity is induced by whatever mechanism or stimulus except for pro-oxidant exposure, BACH1 does not allow NRF2 to act on the clustered-ARE genes. In contrast, when the cell is exposed to pro-oxidants of whatever nature, BACH1 is fast oxidized [19, 49, 86, 93, 96, 97], rapidly detaches from DNA [93, 94, 98] and readily degrades [93, 97].

At the same time, BACH1 is itself an NRF2 target and an extremely interesting one. The BACH1 locus codes for three transcript variants produced as the result of alternative transcription initiation.

Only BACH1 transcript variant 2 is NRF2-dependent: BACH1 intronic +1411 nt ARE is functional and promotes transcription of the second intron of the gene [50]. The authors of this discovery, Jyrkkänen et al., proposed the existence of the BACH1 feed-back circuit along with their discovery of the functional intronic ARE [50].

The existence of the BACH1 feed-back circuit predisposes different expression dynamics of NRF2 targets depending on their ARE structure: once the NRF2 pathway normalizes oxidative status of the cell, newly synthesized BACH1 protein successfully outcompetes NRF2 on the clustered-ARE loci, while allowing it to act further on nonclustered ARE genes (Figure 10).

Figure 10.

The BACH1 feed-back circuit.

In one of our studies, we appeared to observe these events after we released the JUN-imposed suppression of NRF2 in cells treated with hydrogen peroxide: in contrast non-clustered ARE genes (FTH1, CBR3, SQSTM1), HMOX1 expression did not rise—probably because BACH1, being a non-clustered ARE gene, had increased expression in these those settings [34].

3.6. The 26S proteasome/NRF2 feed-back circuit Once polyubiquitinated, NRF2 is degraded by 26S proteasome [86].

Proteasome inhibitors suppress the degradation of NRF2 and stimulate its nuclear import [99].

At the same time, as shown in mice, Nrf2 controls at least eight genes of proteasomal proteins: Psma1, Psma4, Psmb3, Psmb5, Psmb6, Psmc1, Psmc3 and Psmd14 (Figure 11) [2].

Figure 11.

The 26S proteasome/NRF2 feed-back circuit.

To date of preparation of the chapter, no similar results obtained from studies on human cells have been published. Nevertheless, considering the biological significance of the 26S proteasome, it is not

an unfair assumption that the same genes are probably controlled by NRF2 in humans.

Notably, even in human cells, AP-1 controls expression of interferon gamma [100], which, in turn, transactivates PSME1 [101] and PSME2 [102] subunit genes of the 26S proteasome.

Thus, NRF2 pathway activation facilitates NRF2 proteasomal degradation once KEAP1 is re-enabled to sequester NRF2. At least, this system works in mice, and probably in humans, too, especially since two Psme genes are under the NRF2/AP-1 pathway control.

3.7. The MIR-144 feed-back circuit One of numerous NRF2-antagonizing miRNAs, MIR-144, is expressed under control of AP-1 [103]. This miRNA not merely blocks translation of NRF2, but rather degrades the NRF2 mRNA [104, 105].

3.8. The mitochondrial bidirectional circuit Being regulated by reactive oxygen species and signaling background of the cell, the NRF2/AP-1 pathway depends on mitochondrial function, including their vast signaling activity [106]. Surprisingly, the NRF2/AP-1 pathway has been found to modulate mitochondrial biogenesis by two mechanisms. The first one has direct experimental evidences: NRF2 is a transcriptional regulator of NRF1—nuclear respiratory factor (which is not to be confused with NFE2L1, an NRF2 homolog, also known and referred to as NRF1 in this chapter) [107]. NRF1, in turn, controls a battery of nuclear genes required for mitochondrial biogenesis and function [107]. The second mechanism has not been observed in direct experiments and is related to the fact that the NRF1 protein is also

controlled by estrogen receptors—a distinct family of transcription factors [108]. However, it is well-known that estrogen receptors act in an intimate collaboration with AP-1 proteins [109, 110]. For example, FOS alone is required for expression of 37% of estrogen receptors target genes [111]. Even though the mitochondrial biogenesis-related targets of the estrogen receptors may not fall into the AP-1-dependent category, which is to be addressed in the future, there is another doubtless interaction between the estrogen receptors and the NRF2/AP-1 pathway. The already discussed NRF2/AP-1 target, TXNRD1 is a strong promoter of transactivatory function of the estrogen receptors [111], thus the NRF2/AP-1 pathway positively regulates estrogen receptors function, while estrogen receptors facilitate mitochondria biogenesis.

Although mitochondrial biogenesis eventually leads to increased ROS generation, this circuit is unlikely to be purely or even mostly feedforward. Rather, this circuit is bidirectional with strong feed-back action because the outer mitochondrial membrane carries the PGAM5 protein. PGAM5, in turn, tethers the NRF2-KEAP1 complexes to mitochondria and acts as a powerful suppressor of the NRF2dependent expression (Figure 12) [106, 112].

Figure 12.

The mitochondrial bi-directional circuit. NRF1 here is nuclear respiratory factor 1.

Thus, the more mitochondria, the more KEAP1-NRF2 complexes are likely to be bound to them, leading to decreased availability of NRF2 to nucleus. Unfortunately, it is not known to date how PGAM5 expression is controlled with relation to mitochondria biogenesis—the protein is poorly studied.

It is worth noting that the PGAM5/KEAP1-NRF2 system is suggested to represent a stand-by alarm inducible by changes in mitochondrial function [106, 112]. PGAM5 also appears to stimulate mitophagy as a part of cellular defense against excessive reactive oxygen species [113]. It could be suggested that this function of PGAM5 is directly related to its sequestering of NRF2 with subsequent release of it upon mitochondrial dysfunction: NRF2 controls expression

of SQSTM1 and SESN2, two factors of autophagy/mitophagy already discussed above [32, 114].

3.9. The NRF2 bidirectional circuit NRF2 expression is known to respond to stimuli that activate the NRF2/AP-1 pathway [34, 51, 115]. We and others observed that, for reasons still to be revealed, NRF2 had a humped expression curve when different cells had been treated with various pro-oxidant substances [34, 115]. It is possible that this is related to the AP1/TXN/NRF2 bidirectional circuit. It is tempting to suggest that this is caused by AP-1 components that are expressed with different dynamics upon stimulation.

For example, Siriani et al. demonstrated, that angiotensin II, over the same period of treatment, induced 300-fold change in expression of FOS, 500-fold change in expression of FOSB, 2-fold change in expression of FOSL1 and FOSL2, 4-fold change in expression of JUN, 14-fold change in expression of JUNB and 2-fold change in expression of JUND. The same was also fair for the classical AP-1 inducer—TPA: it caused 194-fold change in expression of FOS, 65-fold change in expression of FOSB, 2-fold change in expression of FOSL1, 2-fold change in expression of JUN and JUND, and 9-fold change in expression of JUNB [116].

Since overflow inhibition is characteristic of AP-1 [117], and competition of bZip dimers is a must, it is possible that alterations in ratio of AP-1 components controlling NRF2 expression lead to its humped dosage-expression curve.

This circuit cannot be considered purely feed-back, because before certain threshold, the stimulus activates NRF2 expression. Yet, this character of expression of NRF2 is not entirely of feed-forward nature, and this is discussed in the next, last, section of this chapter.

3.10. The truncated NRF2 feed-back circuit NRF2 is cleaved into a truncated form named truncated NRF2, or tNRF2, by Ice family-like caspases [118]. Just as the full-length protein, tNRF2 enters nucleus, but there it antagonizes the normal NRF2 and suppresses ARE-containing loci [19, 118].

4. Conclusion Although studying the NRF2/AP-1 pathway targets is an extremely complex task due to their plentitude, experimental evidences suggest that this relatively small, with respect to the regulatory proteins, pathway has enormous number of feed-forward and feed-back circuits, some of them being bidirectional.

Some evidences demonstrated directly that these circuits contribute to either fast activation or quenching of the pathway. For other circuits, no such testing has ever been performed. Nevertheless, the existing data unambiguously point out that these circuits are an essential feature of the NRF2/AP-1 pathway. These circuits must be accounted in all applications—starting from cell sensor-based pharmacological screening techniques [119, 120] and ending with therapeutical research and development [121].

https://www.youtube.com/watch?v=Y2Tvy9qKwao

https://www.youtube.com/watch?v=LuUM5Zxe6_A

pequeño circuito protector contra sobre corriente o contra cortos circuitos para salida de voltaje para fuentes de alimentación DC ( OCP ) " fusible electrónico " hola que tal a todos hoy les presento un pequeño circuito el cual nos resulta muy útil cuando trabajamos con fuentes de alimentación y necesitamos protegerlas contra cortos circuitos o por sobre corriente ya sea para una fuente laboratorio o una fuente conmutada casera o una fuente lineal casera trata de un circuito de protección contra cortos circuitos o sobre corriente o fusible electrónico o como se dice abreviado OCP " over current protection " el cual es muy bueno bueno el circuito circuito es muy sencillo y fácil de realizar y utiliza muy pocos en mi caso todos fueron reciclados los materiales que necesitamos para real izarlo son : 1. un diodo in40007 2. un tiristor bt151 o cualquier tiristor similar funcionara muy bien en mi caso e usado un pcr406 3. un transistor s8550 o cualquier pequeño transistor PNP funcionara muy bien como un a733 4. un pulsador normalmente cerrado 5. un relet de 5 pines y de 12 voltio

6.

7.

8.

9.

10.

11. "

un capacitor de 100nf una resistencia de 33k una resistencia de 2.5k una resistencia de 2.2k un led de cualquier color una resistencia de bajo valor " leer mas adelante en el texto

el diagrama para este pequeño circuito protector es el siguiente :

como puede ver el circuito es muy simple y fácil de realizar funciona

muy bien para fuente de alimentación con un voltaje entre 12 voltio a 14 voltio DC y con una corriente máxima que dependerá de la resistencia de bajo valor que mostrare como se calcula mas adelante bueno el circuito funciona de la siguiente manera cuando el circuito esta bien que no tiene carga ni sobre consumo el relet esta apagado pero deja pasar el voltaje por sus pines que están normalmente cerrado ahora cuando hacemos un corto o nos pasamos de la corriente limite el circuito activa el relet separando los pines del normalmente cerrado a normalmente abierto protegiendo la salida de la fuente y la carga en caso de que el corto se haya producido por un accidente bueno el circuito detecta el corto o la sobre corriente utilizando una resistencia de bajo volar que actúa como una resistencia shunf cuando en esa resistencia cae 0.5 voltio el transistor s8550 se activa y a la ves activa el tiristor SCR alimentando el relet y separando sus contactos y cortando el voltaje y la corriente hasta que se reinicie por medio del pulsador normalmente cerrado cuando pulsamos el pulsador normalmente cerrado y todavía se encuentra el corto o la sobre corriente a la salida este se vuelve a proteger inmediatamente la resistencia de bajo valor se calcula con la siguiente formula :

donde R? es el valor de resistencia de bajo a valor a usar si por ejemplo usamos una resistencia de 0.47 ohm aplicando la formula daría el siguiente resultado 0.5 que es el voltaje al cual conduce el transistor entre 0.47 ohm que es el valor de resistencia que hemos elegido es igual a 1 amperio osea que cuando le exijamos mas de un amperio a la fuente esta se protege inmediatamente o si provocamos un corto a la salida ya que cuando se produce un corto a la salida ya que de la fuente circulan varios amperios de ella cosa que la podría dañar pero con este circuito no pasaría esto porque se activaría el circuito protector protegiendo nuestra fuente

fotos del pequeño circuito protector para fuentes de alimentación " OCP " :

en mi caso e usado una protoboar para montar y probar todo le circuito se que me a quedado todo un desastre pero funciona excelente el circuito

como ven aquí ya tenia todo el circuito montado en la protobar para realizar las pruebas de funcionamiento del circuito se que me a quedado todo un desastre pero funciona muy bien :)

prueba de funcionamiento del pequeño circuito " OCP "

aquí ya lo e conectado a mi pequeña fuente de 12 voltio que utilizo mucho para probar mis circuito y se puede ver el led verde encendido indicando que no hay problemas de sobre corriente o de corto circuito a la salida

aquí en estas ultimas fotos se puede apreciar que el led rojo esta encendido indicando que ubo un problema que pudo a ver sido un corto a la salida o un sobre consumo en el circuito y a cortado la corriente de salida y no habrá corriente de salida hasta que se reinicie

el pequeño circuito protector mediante el pulsador normalmente cerrado prueba de funcionamiento del pequeño circuito protector " OCP "

como pueden ver el circuito funciona correctamente en este caso había calculado la resistencia de bajo valor para que el circuito cortara a 2.5 amperios para las pruebas y se puede ver que tenia como carga una pequeña lampara inca decente que cuando yo hago el corto a propósito con el cable el circuito protector se activa cortando la corriente de salida y por consiguiente se apaga la lampara hasta que la reinicie manualmente con el pulsador normalmente cerrado después mas adelante les traeré uno que se pueda regular la corriente máxima mediante un potenciometro o un reotato para que sea mas preciso y usando un opam

CIRCITO BÁSICO OCP

CIRCITO BÁSICO OCP MODIFICADO

ELECTRÓNICA DE POTENCIA/ ELECTRÓNICA INDUSTRIAL 1 Y LABORATORIO, 2547820/2507385 Trabajo grupal 1,10%, carga r, control por fase, disparo aislado de scr,, entrada de consignas, tiempo ida, tiempo regreso por teclado de pc. septiembre 22 inicio.

1.

Trabajo grupal 2,10%, accionamiento completo, motor BALDOR, DC, excitación independiente, técnica PWM, ingreso de consignas desde plataforma computacional. Entrega

2.

Trabajo grupal 3,10%. Funcionamiento, topología, cálculo,, utilidad fr protecciones: OVP para voltaje de salida variable del tipo CROWBAR , OCP tipo FEEDFORWARD en fuente DC, OCP del tipo FEEDBACK o FOLDBACK en fuente DC, OCP del tipo SHUTDOWNP Con sensado diferencial de corriente de entrada.)

3.

Eamen grupal #1 /2020-2/15% 4. Eamen grupal #2 /2020-2/15%.

5.

Simulación grupal, SPICE circuito, 15%

6.

Simulación grupal, LABVIEW, formas de 7. Seguimiento y/o emulación de laboratorio,