EE Power & Energy Efficiency Handbook 2016

September 2016

Energy efficiency in LLC resonant conversion topologies Page 22

Better thermal design means better efficiency Page 37

Power & Energy Efficiency H A NDBO O K

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Power & Energy Efficiency

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what’s inside 37

26

40 06

Energy efficient doesn’t mean cost effiicient

08

A better way to squelch transients

15

Modern dc-dc converters can double as compact transient suppressors in vehicular electrical systems characterized by a lot of hash on the battery power lines.

Choosing inductors for energy efficient power applications Energy efficiency in power supplies can be as much about the inductors as about the circuit topology. A few simple calculations can prevent unpleasant surprises.

22

Energy efficiency in LLC resonant conversion topologies

Designers are now investigating exotic power supply topologies in a quest for efficient operation.

26

DC-DC converters with renewable energy in mind

Modern battery, fuel cell and solar cell systems need ways of transferring power in two directions. New dc-dc converters that are bidirectional can eliminate the need for bulky transformers in these applications while bringing more efficient operation.

32

Bumpless control for power factor correction

It can be difficult to correct for power factor in converters that have different conduction modes. A special technique can make corrections smoothly.

37

Better thermal design means better efficiency

The strategic use of computational fluid dynamics can speed the development of high-tech products.

40

Customized approach to better efficiency

VPX power systems are a defense industry mainstay. New standards help boost efficiency by eliminating the need to overspecify supplies when handling outlier situations.

44

Guaging energy efficiency in complex motor drives

Modern instrumentation can help reveal sources of power dissipation problems where electric motors are tightly integrated with their controls and test points are hard to find.

C O V ER P H O TO G R A P H Y B Y A L L I S O N WA S H K O

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Power & Energy Efficiency

Energy efficient doesn’t mean cost efficient LEE TESCHLER EXECUTIVE EDITOR

I ONCE

had a root cellar illuminated by a single incandescent light bulb, the kind of bulb that is being phased out thanks to energy efficiency regulations. I would use that light bulb perhaps 15 minutes over the span of a year. When it burned out, DOE regulators would have had me replace it with a compact fluorescent bulb costing about five times as much as the original incandescent. Turned on only 15 minutes/year, there was no way the increased efficiency of the CFL bulb would ever make up for its price tag.

I recalled this incident when reviewing a recent report on whether energy efficiency regulations put in place by the Dept. of Energy are worth what they cost. DOE efficiency regulations now apply to everything from refrigerators and water heaters to lamp ballasts and electric motors. The Office of Information and Regulatory Affairs (OIRA) says these mandates have posed the third-highest cost burden on U.S. citizens from 2002 to 2012 of any regulatory agency, behind only those from the DoT and EPA. Of course, costs are OK if their benefits more than pay for them. DOE’s own estimates are that from 2007 to 2015 its regulations have cost $9.5 billion but have brought $32 billion in benefits. The problem, points out a think tank called the American Action Forum, is that both figures are subject to a large amount of uncertainty. One issue is that they are based on economic models rather than on real data.

The economic models use a discount rate to figure out whether money spent today on more efficient products will be earned back

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from benefits over time because of lower energy use. But scholars from the George Washington Regulatory Studies Center who have looked at DOE regulatory assumptions have found that small adjustments in the discount rates can turn a rule with benefits into something that costs more than it’s worth.

More worrisome, says AAF, there is no such thing as a “peer review” outside the DOE for the assumptions that go into economic models justifying rules. That leads to energy regulations that are on shaky ground economically.

Fortunately, some energy regulations have been in place long enough for scholars to study what actually happens when they’re implemented rather than what models predict. The results aren’t promising.

In one case, National Bureau of Economic Research scholars looked at more than 30,000 households that had participated in a DOE weatherization assistance program.

The researchers found the upfront investment costs to be about twice the actual energy savings.

Big surprise: The economic model DOE used to justify the program turned out to be flawed. It projected savings of about 2.5 times what was actually realized.

Even worse, those who participated in the program didn’t seem to be any warmer in the winter. NBER scholars found no evidence of significantly higher indoor temperatures at weatherized homes. And even when accounting for the broader societal benefits of energy efficiency investments, they say, the costs still substantially outweigh the benefits; the average rate of return is approximately -9.5% annually.

I suspect that roughly the same rate of negative return applies when incandescent bulbs used a few minutes annually get replaced with much more expensive energy efficient versions.

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Power & Energy Efficiency

A better way to squelch transients Modern dc-dc converters can double as compact transient suppressors in vehicular electrical systems characterized by a lot of hash on the battery power lines.

TIMOTHY HEGARTY

NON-ISOLATED POWER SOLUTIONS, TEXAS INSTRUMENTS INC.

THE

proliferation of electronic subsystems in vehicles has created a need for small, inexpensive and highly reliable electronics able to operate in challenging conditions. A number of these conditions arise because of noise on vehicle power rails. A vehicle battery’s steady-state range is 9 to 16 V, depending on its state of charge, temperature, and the condition of the alternator. However, the power rail is also subject to a range of dynamic disturbances, including startstop, cold-crank and load-dump transient profiles.

All these kinds of events create electrical conditions that can be problematic for electronics. To test for vulnerabilities, each vehicle manufacturer has its own extensive conducted immunity (CI) test suite, and there are standardized pulse waveforms given by international standards such as ISO 7637 and ISO 16750. Besides common sources of undervoltage and overvoltage transients, alternator sinusoidalprofiled noise superimposed on the dc bus can be deleterious, particularly for vehicle infotainment and lighting systems.

AUTOMOTIVE POWER LINE TEST LEVELS

Test levels for automotive power line continuous and transient conducted disturbance tests

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Power & Energy Efficiency

ISO 16750-2 superimposed sinusoidal ac test voltage (a), log frequency sweep from 50 Hz to 25 kHz over a two-minute sweep duration (b).

ISO TEST WAVEFORMS

Most vehicles try to headoff transient disturbances using a passive circuit consisting of a low-pass inductor plus capacitor (LC) filter and transient voltage suppressor (TVS) array. Automotive electronics sitting downstream from the protection network are then rated to survive up to a 40-V transient without damage.

But the inductor/electrolytic capacitor combination needed for attenuating disturbances at low frequencies takes up a lot of space.

Fortunately, there is a more compact way of dealing with transients. It involves an active filter using a synchronous buck-

EFFICIENCY vs LOAD CURRENT

boost dc/dc converter having a high power supply rejection ratio (PSRR, where rejection expressed as a log ratio of output noise to input noise). Besides handling filtering, the active filter also provides both battery voltage regulation and transient rejection.

SYNCHRONOUS BUCK-BOOST CONVERTER When devising regulators that work with vehicle batteries, designers must keep in mind that the battery voltage can be above the output voltage setpoint (as when the battery is charging), below the setpoint (as when severely discharging),

and equal to the setpoint. This variability calls for a buck-boost conversion. Traditional buck or boost converters are inadequate here because only a step-down or step-up conversion is possible, respectively.

A four-switch synchronous buck-boost converter can be designed around an LM5175-Q1 controller to output a tightly-regulated 12-V rail. This approach works well for engine management units (EMU) and other critical automotive functions where loads must remain powered without glitches during even the most severe battery-voltage transients.

EFFICIENCY vs INPUT VOLTAGE

Efficiency plot and component power loss breakdown versus (a) load current and (b) input voltage for the four-switch synchronous buck-boost converter. The buck-boost mode region is evident in the efficiency versus line plot.

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TRANSIENTS The main attraction of this modern buck-boost power stage is that simple buck or boost operating modes are used to realize a high conversion efficiency. The circuit produces a positive output voltage in contrast to the classic single-switch (inverting) buck-boost. And, by virtue of its simple magnetic component, it exhibits less power loss and higher power density relative to SEPIC, flyback, Zeta, and cascaded boost-buck topologies.

The four-switch buck-boost converter has an intuitive topology. It is also relatively compact and goes through a controlled startup while incorporating short-circuit protection in boost mode. As well, the control and compensation is simple and the converter uses a constantswitching frequency. As such, this approach works well for automotive battery voltage regulation.

A schematic of a typical four-switch buck-boost converter circuit specifies components for a power stage and a LM5175-Q1 controller chip. The controller chip includes integrated gate drivers, a bias supply,

current sensing circuits, output voltage feedback, loop compensation, a programmable under-voltage lock-out circuit, and a dither option for lower noise signature. The designer can choose a switching frequency. A 400 kHz switching frequency will minimize the circuit footprint and eliminate interference with the AM broadcast band.

In the accompanying schematic, four power MOSFETs are arranged as buck and boost legs in an H-bridge configuration, with switch nodes SW1 and SW2 connected by a power inductor, designated LF. Conventional synchronous buck or boost operation takes place when the input voltage lies suitably above or below the output voltage, respectively, and the high-side MOSFET of the opposite, non-switching leg conducts as a pass device.

However, the most compelling feature of this particular buck-boost implementation is that it employs a special scheme in the buck-boost (B-B) transition region when the input is close to the output voltage setpoint. Then, both buck and boost legs each switch at half the switching

FOUR-SWITCH BUCK-BOOST CONVERTER

This circuit realizes a four-switch buck-boost converter with combined peak-and-valley current-mode control. The circuit produces a conventional synchronous buck or boost operation when the input voltage lies suitably above or below the output voltage, respectively, and the high-side MOSFET of the opposite, non-switching leg conducts as a pass device.

COMPONENT LIST, FOUR-SWITCH CONVERTER

Here are the components used to realize the four-switch synchronous buck-boost converter.

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Power & Energy Efficiency

PHASE vs FREQUENCY

PSRR vs FREQUENCY A simulation of the converter’s openloop gain and phase plots at input voltages of 9 and 16 V shows the corresponding PSRR performance. At left is the simulated bode plots at 9 and 16-V inputs with loop crossover frequencies of 14 kHz and 17 kHz, respectively. At right is a simulated PSRR highlighting attenuation at 1 kHz of 40 dB and 42 dB at 9 V and 16 V inputs, respectively.

frequency in a phase-shifted, interleaved manner that is particularly advantageous for efficiency and power loss.

A control architecture that combines peak current-mode control in boost and valley current-mode control in buck enables smooth mode transitions, requiring just one lowside configured shunt resistor for current sensing.

It is interesting to view plots of efficiency and component power dissipations versus line and load for the accompanying converter design. Considering losses in totality, a converter with 12-V regulated output quite readily hits efficiencies in excess of 95% across wide ranges of output current and input voltage.

One important property for vehicular electronics is immunity to conducted transients in the audio frequency (AF) range. A source of this noise is the automotive alternator which has a residual alternating current on its output. The alternator’s stator winding is basically a threephase sinusoidal current source with high-impedance

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output feeding into a diode full-wave rectifier. The rectification creates overlapping current pulses and the ripple is determined from the three phases.

ISO 16750-2 section 4.4 describes the voltage ripple on the alternator’s output that vehicular electronics should be able to withstand. The test signal is in the frequency range of 50 Hz to 25 kHz with an amplitude of 1 V, 2 V and 4 V peak-to-peak, depending on the test pulse severity.

MAXIMIZING PSRR PSRR of a dc/dc converter is related to and affected by loop bandwidth. The loop bandwidth is typically limited to 20% of the switching frequency or lower, depending on the right-half-plane zero (RHPZ) frequency that appears when operating in boost mode. In a controller such as TI’s LM5175-Q1, PSRR performance is largely independent of VIN and load changes. This is thanks to a current-mode

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Power & Energy Efficiency

control scheme with adaptive slope compensation, based on the difference of VIN and VOUT, that is designed to raise PSRR and reject line transients.

In the accompanying schematic, the capacitor designated CSLOPE establishes the slope compensation setting. CSLOPE is chosen for an idealized deadbeat response at the classic one-times the inductor upslope (downslope) in valley (peak) current-mode buck (boost) mode of operation. A setting of half the applicable inductor slope theoretically provides optimal line rejection, but this represents the minimum slope compensation for loop stability.

All in all, the four-switch synchronous buck-boost converter is a compact and economical way of realizing tight voltage regulation in automotive applications. It obviates the need for bulky passive filter components. The buck-boost controller also has an AEC-Q100 automotive qualification.

REFERENCES

Singh, Atul. “Load dump and cranking protection for automotive backlight LED power supply,” TI application report (SNVA681A), March 2015 www.ti.com/lit/snva681 “ISO 7637-2:2011, Road vehicles – electrical disturbances from conduction and coupling,” ISO standards website: www.iso.org/iso/catalogue_detail.htm?csnumber=50925 “ISO 16750-2:2012, Road vehicles – environmental conditions and testing for electrical and electronic equipment,” ISO standards website: www.iso.org/iso/catalogue_detail.htm?csnumber=61280 “Front end power supply reference design with cold crank operation, transient protection, EMI filter,” TI Design (TIDA-00699) www.ti.com/tool/tida-00699 Hegarty, T. “Optimizing the efficiency of the 4-switch buck-boost converter,” How2Power, September 2015 www.how2power.com/newsletters/1509/articles/ H2PToday1509_design_TexasInstruments.pdf Choudhary, V. and Jacob, M. “Smart diode and 4-switch buck-boost converter provide ultra-high efficiency, compact solution for 12-V automotive battery rail,” PCIM Europe, Nuremburg, May 10–12, 2016 LM5175EVM-HD 400 kHz high density buck-boost converter EVM, Texas Instruments www.ti.com/tool/lm5175evm-hd “Automotive cranking simulator user’s guide,” Texas Instruments (slvu984) www.ti.com/lit/slvu984

At top is the synchronous buck-boost converter’s output voltage when the 9-V dc input has a superimposed 1 kHz sinusoidal ripple of 1-V peak-to-peak amplitude. All voltages were measured with an ac scope probe coupling with the switching frequency noise removed. The input voltage modulation is via a series n-channel MOSFET connected as a source follower. The input signal is attenuated by approximately 40 dB as expected. At left, bottom, is the output voltage during a cold-crank transient down to 3 V for 20 msec. using an automotive cold-crank simulator. As presented, the four-switch synchronous buckboost converter regulates seamlessly through the coldcrank profile, maintaining an output voltage at its nominal 12-V setpoint. The power MOSFETs have adequate gate drive amplitude at low input voltage as VOUT powers the controller’s BIAS pin input.

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CHOOSING INDUCTORS

Choosing inductors for energy efficient power applications CHRIS HARE, LEONARD CRANE COILCRAFT, INC.

Energy efficiency in power supplies can be as much about the inductors as about the circuit topology. A few simple calculations can prevent unpleasant surprises.

IN

high frequency dc-dc converters, inductors filter out the ac ripple current superimposed on the dc output. Whether the converter steps the voltage down – buck – or steps the voltage up – boost – or both up and down – SEPIC - the inductor smooths the ripple to provide a pseudo-dc output.

Applications with batteries can extend battery life by improving the efficiency of the entire power supply circuit, and inductor efficiency is often a major part of the design. Careful consideration of inductor efficiency can mean the difference between having your battery work when you need it and having it die in the middle of an important task.

Inductor efficiency is highest when the combination of core and winding losses are at a minimum. Therefore, highest efficiency comes from selecting an inductor that provides sufficient inductance to smooth out the ripple current while simultaneously minimizing losses. The inductor must pass the current without saturating the core or over-heating the winding.

It can be fairly complicated to accurately predict an inductor’s core and winding loss. Core loss depends on several factors, such as peak-peak ripple current, ripple current frequency, core material, core size, and turn count. The required ripple current and ripple current frequency are application-dependent, while the core material, core size, and turn count are inductor-dependent.

The most commonly-used equation to characterize core loss is the Steinmetz equation: x

Pcore = K(f) (B)

y

Where Pcore = power loss in the core, W; K, x, y = core material constants; f = frequency, Hz; and B = flux density, T. This equation shows that core loss depends on frequency f and flux density B. Flux density depends on ripple current, so both are applicationdependent variables. The equation also shows that the core loss is inductor-dependent, where the core material determines the K,

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Coilcraft_P&EE8-16_V5.indd 15

x, and y constants. Note flux density is also a function of the core area (Ae) and the number of turns (N), so core loss is both applicationdependent and inductor-dependent.

By comparison, dc winding loss is simple to calculate: Pdc = ldc2 × Rdc Where Pdc = dc power dissipated, W; Idc = effective dc (rms) value of the inductor current, A; and Rdc = dc resistance of the inductor winding, Ω.

AC winding loss is more complicated and may include the effects of increased resistance at higher frequency caused by both skin and proximity effects. ESR (effective series resistance) or ACR (AC resistance) curves may show some of the increased resistance at higher frequency.

However, these curves are typically made at low current levels, so they do not capture current-dependent (core) loss. They are also subject to possible misinterpretation.

For example, consider the ESR vs frequency curve in the accompanying figure.

An initial observation is that the resistance looks high above 1 MHz. This behavior would strongly suggest that the inductor not be used at that frequency because of the high loss from ESR. However, it has been observed that parts with curves like this have performed well in actual converters – much better than the curves would suggest.

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1000

100

We can calculate the loss in the inductor: l2R = (0.3)2 × (0.8) = 0.072 0.072 ÷ 1.5 = 0.048 ~ 5% Thus approximately 5% of output power lost in the inductor. However, if we were to run the same converter at 5 MHz, we can see from the ESR curve that R is between 10 and 20 Ω. If we assume R = 10 Ω, the power loss in the inductor should be: l2R = (0.3)2 × (10) = 0.9 0.9 ÷ 1.5 = 0.60

Equivalent SeriesResistance Resistance (Ohms) Equivalent Series (Ohms)

Power & Energy Efficiency

WINDING ESR vs FREQUENCY

Consider the following example: Assume a converter must provide an output of 5 V at 0.3 A (1.5 W). We use a 10-µH Coilcraft inductor with a typical ESR vs frequency as in the accompanying figure. If the converter operates at 250 kHz, the graph shows that the ESR, which includes both ac and dc resistance, is approximately 0.8 Ω.

For a buck converter, the average inductor current equals the load current, 0.3 A.

10

1

0.1

10 µH

0.01

0.1

1

Frequency Frequency (MHz) (MHz)

10

A quick look at the ESR vs frequency curve shows that the resistance looks high above 1 MHz. This would strongly suggest that this part cannot or should not be used at that frequency because of the high loss from the ESR. However, it has been observed that parts with curves like this have performed well in actual converters – much better than would be suggested by these curves.

THE DC-DC CONVERTER INDUCTOR SELECTOR For dc-dc converters, the Coilcraft dc-dc converter inductor selector calculates the inductance value, peak current, and peak-peak current requirements based on operating conditions and amount of ac ripple current chosen. It then feeds this information into a Power Inductor Finder tool to display a list of inductors that may meet these requirements.

The list includes the inductance at peak current, current rating, total losses, and resulting part temperature for each inductor listed.

Inductor loss is closely related to core size and wire size.

In many cases, lowest loss corresponds to larger part size, or it corresponds to using a hard-saturation core material. As with any design, there may be compromises that require analyzing trade-offs in size or inductance at peak current vs efficiency. Having all of the inductor information in a complete list that allows multiple sorting facilitates such an analysis.

If you already know the inductance value and current ratings required for your application, you can enter this information directly into the Power Inductor Finder. The results include core and winding (total) loss and saturation current ratings for each inductor, to verify that the inductance will remain close to the design requirement at the peak current condition for your application.

The tool may also be used to graph the inductance-vs-current behavior to compare traditional hard-saturating inductors to soft saturation types. To select the highest efficiency inductor, the results can be first sorted by total loss.

Multiple sorts allow selection by multiple parameters.

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Power & Energy Efficiency

Converter current

ton toff

Continuous Current Continuous Mode Current Mode

Time A simplified version of a possible buck converter waveform with continuous conduction shows a ripple current that is relatively small compared to the average current.

TM TM 850VUltra-Junction Ultra-JunctionX-Class X-ClassHiPerFET HiPerFETTM PowerMOSFETs MOSFETs 850V Ultra-Junction X-Class HiPerFET Power MOSFETs 850V TMPower 850V Ultra-Junction X-Class HiPerFET Power MOSFETs Enablingvery veryhigh highpower powerdensity density- -industry's industry'slowest loweston-resistance on-resistanceSilicon Silicon devices Enabling devices

Enabling very high power density - industry's lowest on-resistance Silicon devices Enabling very high power densityFeatures: - industry's lowest on-resistance Silicon devices Features: Advantages: Applications: Features: Advantages: Applications: Advantages: Applications: TO-220 TO-220 TO-220

TO-263HV TO-263HV TO-263HV

SOT-227 SOT-227 SOT-227

TO-263HV

SOT-227

S

TO-247 TO-247 TO-247 TO-247

SS

TO-268HV TO-268HV TO-268HV

S

D

DD

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TO-220

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TO-268HV

PLUS264 PLUS264 PLUS264

TO-264

PLUS264

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Part Part Part Number Number Number Part Number IXFH20N85X IXFH20N85X IXFH20N85X IXFP20N85X IXFP20N85X IXFP20N85X IXFH20N85X IXFA20N85XHV IXFA20N85XHV IXFA20N85XHV IXFP20N85X IXFH40N85X IXFH40N85X IXFH40N85X IXFA20N85XHV IXFT40N85XHV IXFT40N85XHV IXFT40N85XHV IXFH40N85X IXFT50N85XHV IXFT50N85XHV IXFT50N85XHV IXFT40N85XHV IXFK50N85X IXFK50N85X IXFK50N85X IXFT50N85XHV IXFH50N85X IXFH50N85X IXFH50N85X IXFK50N85X IXFN66N85X IXFN66N85X IXFN66N85X IXFH50N85X IXFX66N85X IXFX66N85X IXFX66N85X IXFN66N85X IXFK66N85X IXFK66N85X IXFK66N85X IXFX66N85X IXFB90N85X IXFB90N85X IXFB90N85X IXFK66N85X IXFN90N85X IXFN90N85X IXFN90N85X IXFB90N85X IXFN110N85X IXFN110N85X IXFN110N85X IXFN90N85X IXFN110N85X Features:

VDSSVDSS VDSS Max Max MaxVDSS (V)(V) (V)Max 850850 850 (V) 850850 850850 850850 850850 850850 850850 850850 850850 850850 850850 850850 850850 850850 850850 850850 850850 850850 850850 850850 850850 850850 850850 850850 850850 850850 850850 850

•Advantages: Higher efficiency Higher efficiency •• Higher efficiency • High power density • High power density • Higher efficiency • High power density ••Easy Easy to mount tomount mount High power density •• Easy to ••Space Space savings Easysavings tosavings mount •• Space • Space savings

RDS(on) RDS(on) RDS(on) I ID(cont) I D(cont) Max Max Max RDS(on) = 25°C TCD(cont) =TC25°C TC = 25°C I TJ=25°C TJ=25°C TJ=25°C Max (A) (A)D(cont) (A) TC = 25°C (Ω)(Ω) (Ω) TJ=25°C 20 0.33 20(A) 0.33 20 0.33(Ω) 0.33 0.33 20 20 20 20 0.330.33 0.33 20 20 0.33 0.330.33 20 20 0.145 40 40 0.145 40 20 0.145 0.33 0.145 40 40 0.145 40 40 0.145 0.145 0.105 0.105 50 50 50 40 0.105 0.145 0.105 50 50 0.105 50 50 0.105 0.105 0.105 50 50 0.105 50 50 0.105 0.105 0.065 65 65 0.065 0.065 65 50 0.105 0.065 66 66 0.065 66 65 0.065 0.065 0.065 66 66 0.065 66 66 0.065 0.065 0.041 90 90 0.041 90 66 0.041 0.065 0.041 90 90 0.041 0.041 90 90 0.041 0.033 110110 0.033 0.033 110 90 0.041 110 Advantages: 0.033

QQ Qg g g TypTyp Typ Qg (nC) (nC) (nC)Typ 63 63(nC) 63 63 63 63 63 63 63 63 63 98 98 98 63 98 98 98 98 152152 152 98 152152 152152 152152 152152 230230 230152 230230 230230 230230 230230 340340 340230 340340 340340 425425 425340 425

•Applications: Industrial switched-mode and Industrial switched-mode and •• Industrial switched-mode and resonant-mode power supplies resonant-mode power supplies • Industrial switched-mode and resonant-mode power supplies •Electric Electric vehicle battery chargers vehicle battery chargers resonant-mode power supplies •• Electric vehicle battery chargers ••ACAC and motor driveschargers and DCDC motor drives Electric vehicle battery •• AC and DC motor drives • DC-DC converters • DC-DC converters • AC and DC motor • DC-DC converters drives ••Renewable-energy Renewable-energy inverters inverters DC-DC converters •• Renewable-energy inverters ••Power Power Factor Correction (PFC) circuits Factor Correction (PFC) circuits Renewable-energy inverters •• Power Factor Correction (PFC) circuits • Robotics and servo control • Robotics and servo control • Power Factor Correction (PFC) circuits • Robotics and servo control • Robotics and servo control RthJC t t RthJC trrrr rr RthJC Max TypTyp Max Typ trr MaxRthJC (ns) (°C/W) (ns) (°C/W) (ns)Typ (°C/W) Max 0.23 190 190(ns) 0.23 (°C/W) 190 0.23 0.23 190190 0.23 190190 0.230.23 190 0.23 0.23 190 190190 0.230.23 0.145 200200 0.145 200190 0.145 0.23 0.145 200200 0.145 200200 0.145 0.145 0.14 218218 0.14 0.14 218200 0.145 0.14 218218 0.14 218218 0.140.14 0.14 218218 0.14 218218 0.140.14 0.15 250250 0.15 0.150.14 250218 0.10.1 250250 250250 0.10.15 250250 0.10.1 250250 0.1 0.1 0.07 250250 0.07 250250 0.070.1 0.104 250250 0.104 250250 0.104 0.07 0.107 205205 0.107 205250 0.107 0.104 0.107 205 Applications:

Package Package Package Type Type Type Package Type TO-247 TO-247 TO-247 TO-220 TO-220 TO-220 TO-247 TO-263HV TO-263HV TO-263HV TO-220 TO-247 TO-247 TO-247 TO-263HV TO-268HV TO-268HV TO-268HV TO-247 TO-268HV TO-268HV TO-268HV TO-268HV TO-264P TO-264P TO-264P TO-268HV TO-247 TO-247 TO-247 TO-264P SOT-227 SOT-227 SOT-227 TO-247 PLUS247 PLUS247 PLUS247 SOT-227 TO-264P TO-264P TO-264P PLUS247 PLUS264 PLUS264 PLUS264 TO-264P SOT-227 SOT-227 SOT-227 PLUS264 SOT-227 SOT-227 SOT-227 SOT-227 SOT-227

850V Ultra-Junction X-Class HiPerFETTM Power MOSFETs Enabling very high power density - industry's lowest on-resistance Silicon devices TO-263HV SOT-227

uits s

TO-247

PLUS264

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TO-264

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Coilcraft_P&EE8-16_V5.indd 18

uits s

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TO-268HV

EUROPE USA ASIA EUROPE USA ASIA EUROPE ASIA •IXYS Ultra low on-resistance RUSA efficiency • Korea Industrial switched-mode and GmbH IXYS Power• Higher IXYS IXYS Taiwan/IXYS IXYS GmbH IXYS Power IXYS Taiwan/IXYS Korea DS(ON) EUROPE USA ASIA IXYS GmbH IXYS Power Taiwan/IXYS Korea and gate charge Qg • High power density resonant-mode power supplies marcom@ixys.de sales@ixys.com sales@ixys.com.tw marcom@ixys.de sales@ixys.com sales@ixys.com.tw IXYS GmbH IXYS Power IXYS Taiwan/IXYS Korea marcom@ixys.de sales@ixys.com •+49 Fast body diode • Easy tosales@ixys.com.tw mount • Electric vehicle battery chargers 6206-503-249 408-457-9042 sales@ixyskorea.com +49 (0)(0) 6206-503-249 +1+1 408-457-9042 sales@ixyskorea.com marcom@ixys.de sales@ixys.com sales@ixys.com.tw +49 (0) 6206-503-249 +1 408-457-9042 sales@ixyskorea.com • dv/dt ruggedness • Space savings • AC and DC motor drives +49 (0) 6206-503-249 +1 408-457-9042 sales@ixyskorea.com • Avalanche rated • DC-DC converters www.ixys.com www.ixys.com www.ixys.com • Low package inductance • Renewable-energy inverters www.ixys.com • International standard packages • Power Factor Correction (PFC) circuits • Robotics and servo control Part Number

VDSS Max (V)

ID(cont) TC = 25°C (A)

RDS(on) Max TJ=25°C (Ω)

Qg Typ (nC)

trr Typ (ns)

s

TO-220

D

es s

•Features: Ultra low on-resistance RDS(ON) Ultra low on-resistance DS(ON) •• Ultra low on-resistance RRDS(ON) and gate charge Q and gate charge Q g g • Ultra low on-resistance RDS(ON) and gate charge Qg •Fast Fast body diode Q body diode and gate charge •• Fast body diode g ••dv/dt dv/dt ruggedness ruggedness Fastruggedness body diode •• dv/dt • Avalanche rated • Avalanche rated • dv/dt • Avalancheruggedness rated ••Low Low package inductance package inductance Avalanche rated •• Low package inductance ••International International standard packages standard packages Low package inductance •• International standard packages • International standard packages

G TO-264 TO-264 TO-264

V

ss

Ip-p

Idc

G

es s

Buck Converter Buck Converter Inductor Current Inductor Current

GG

ss

CONVERTER CURRENT

Thus 60% of the output power is lost in the inductor! Based on this simple example, it would seem obvious that a designer should not choose a component like this.

It has been observed that converters, in fact, often realize better performance than the ESR curves predict. The following explanation illustrates why.

The nearby figure shows a simplified version of a possible buck converter waveform, with continuous conduction. The ripple current is relatively small compared to the average current.

Assume the ripple current peak-peak is about 10% of the average current. From the previous example this means Idc = 0.3 A and Ip-p = 0.03 A. To predict the inductor losses correctly, current must be separated into two components. For the low-frequency or dc loss, we use the lowfrequency resistance (effectively Rdc), which we can see from the graph is 0.7 Ω. The current is the rms value of the load current plus the ripple current. In this case the ripple current is small, so the value is approximately equal to the dc load current.

RthJC Max (°C/W)

Package 9/29/16 2:37 PM Type

CHOOSING INDUCTORS

So, the total inductor loss at 5 MHz is 0.063 W + 0.00076 W = 0.06376 W. This loss is more significant, about 1.2% greater than Rdc loss, but is not nearly the 0.9 W originally predicted by multiplying the ESR by the entire load current. Also, this example is not exactly fair, because we wouldn’t use the same inductor value at 5 MHz as we would at 250 kHz. We would use a much smaller L and therefore we would get a much smaller Rdc. In summary, the inductor loss must be calculated by a combination of the Rdc and ESR. Losses will be reasonable for a continuous current mode converter in which the ripple current is small compared to the load current.

In typical applications, ripple current is kept to approximately 40% of the load current or less. Regardless of ripple content, ESR curves do not capture current-dependent core loss at higher current. And total inductor loss determines the overall inductor efficiency.

Therefore, inductor manufacturers optimize inductor efficiency by selecting low-loss materials and designing inductors for minimal total loss. The use of rectangular “flat” wire may provide the lowest Rdc in a given size to minimize dc loss. Improvements in core materials have led to inductors with low ac core loss at high frequency, resulting in higher inductor efficiency.

For example, Coilcraft has just released the XEL series of molded power inductors that are optimized for high-frequency,

Low-frequency loss = ldc2R = (0.3)2 × (0.7) = 0.063 W To get the total loss, we must add low-frequency loss to the high-frequency loss, which is I2R. In this case the R is the ESR and the I is the rms value of the ripple current only.

Approximate rms ripple current = lp-p ÷ 2√3 = 0.03 ÷ 3.464 = 0.0087 A At 250 kHz the ac loss would be: I2R = (0.0087 A)2 × (0.8 Ω) = 0.00006 W. Therefore, at 250 kHz, we predict the total inductor loss is 0.063 W + 0.00006 W = 0.06306 W. We see that operating at 250 kHz predicts only slightly more loss (less than 1%) than predicted simply

by Rdc. Now, let’s look at the same example at 5 MHz. The low frequency loss is still the same 0.063 W. The ac loss calculation must use the ESR, which was previously estimated at 10 Ω: I2R = (0.0087 A)2 × (10 Ω) = 0.00076 W.

ss Ts s

SELECTED INDUCTOR QUALITIES

s es

ts s cuits

L nom

DCR typ

Isat (30%)

XEL4020-222

2.2 µH

35.2 mOhms

5.9 A

XAL4020-222

2.2 µH

35.2 mOhms

5.6 A

XFL4020-222

2.2 µH

21.4 mOhms

3.7 A

INDUCTER vs. CURRENT 4.0

XEL4020-222 XAL4020-222 XFL4020-222

7 0 HV 7 HV HV P 7 7 7 P 4 7 7

s

s

ts

Inductance (µH) Inductance (�H)

e

3.0 2.0 1.0 0

0

1

2

3

4

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6

Current Current (A) (A)

7

8

9

10

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The inductance vs current curves for the 2.2 µH value in the XEL, XAL, and XFL inductor series from Coilcraft.

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TOTAL LOSSES VS. PEAK TO PEAK RIPPLE CURRENT

(2.2�H AT 3 Current MHz) Total Losses vs. Peak to Peak Ripple (2.2 µH at 3 MHz) 0.6

XEL4020-222 XAL4020-222 XFL4020-222

TotalLosses Losses (W)(W) Total

0.5 0.4 0.3 0.2 0.1 0

0

0.2

0.4

0.6

0.8

Peak to to Peak Current Peak PeakRipple Ripple Current (A) (A)

1.0

1.2

AC LOSS VS. PEAK TO PEAK RIPPLE CURRENT (2.2�H AT

3 MHz) AC Loss vs. Peak to Peak Ripple Current (2.2 µH at 3 MHz) 0.8

AC (W) AC Loss Loss (W)

Power & Energy Efficiency

high-peak-current applications. These inductors are designed for soft-saturation, while giving the lowest ac loss at frequencies of 2 MHz and higher.

The nearby graph of inductance vs current characteristics is for the 2.2 µH value in the XEL, XAL, and XFL series.

The XEL and XAL series are clearly the best choice for holding inductance at currents of around 3 A or higher. Another nearby graph compares the ac loss and total loss of the same inductors at 3 MHz. Although the XFL inductor has the lowest total power loss, the new XEL inductor has lower total loss than the XAL and is therefore the best choice for highfrequency power converter applications that must withstand high peak current.

To speed the design process for engineers selecting inductors, Coilcraft has developed tools that calculate measurement-based core and winding loss for each possible application condition. The results from these tools include current-dependent and frequency-dependent core and winding loss, eliminating the need for requesting proprietary inductor design information, such as core material, Ae, and number of turns, and the need to perform hand calculations.

Designing for highest efficiency requires selection of inductors with the lowest total loss at application conditions.

Calculating total loss can be complicated, but these calculations are built into Coilcraft power magnetics tools, making selection, comparison, and analysis as simple as possible.

XEL4020-222 XAL4020-222 XFL4020-222

0.6

40% improvement

0.4 0.2 0

0

0.2

0.4

0.6

0.8

1.0

1.2

1.4

1.6

Peak to Peak Ripple Current (A)

1.8

2.0

AC loss and total loss of XEL, XAL, and XFL inductor series from Coilcraft at 3 MHz. Although the XFL inductor has the lowest total power loss, the XEL inductor has lower total loss than the XAL and is therefore the best choice for high frequency power converter applications that must withstand high peak current.

REFERENCES Inductor Performance in High Frequency dc-dc Converters, Len Crane, Document 470-2, 2015 www.coilcraft.com/pdfs/Doc470_Inductor_performance_in_dc-dc_converters.pdf XEL, XAL, or XFL? – Making the Best Choice www. coilcraft.com/xal_or_xfl.cfm Coilcraft DC-DC Inductor Selector, Power Magnetics Tools www. coilcraft.com/apps/selector/selector_1.cfm Coilcraft Power Inductor Finder, Power Magnetics Tools www. coilcraft.com/apps/finder/finder.cfm

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Power & Energy Efficiency

Energy efficiency in LLC Resonant Conversion Topologies RICK BARNETT, ED FONTANA, JIM MONTGOMERY, EMBEDDED SOLUTIONS, GE INDUSTRIAL SOLUTIONS BUSINESS

Designers are now investigating exotic power supply topologies in a quest for efficient operation.

IT’S

no secret that power converters increasingly emphasize energy efficiency. For example, 115-Vac computer power supplies must exhibit a 94% energy efficiency at 50% of their rated load to receive an 80 Plus Titanium rating; 230-Vac supplies must hit 96% at half their rated load to get the designation.

For external power supplies (as used by laptops, tablets, and so forth), the Dept. of Energy recently came out with Level VI energy efficiency standards mandating active-mode efficiencies in the 86 to 88% range from most output powers.

A generally understood maxim is that for power supplies to get denser, they must also get more efficient. That works up to a point. Then you reach a point where incremental increases in efficiency come only at great cost and effort, or alternately, you must give relief on one to get the other.

As a generality, there's a brick wall in 48-V power at around 50 W/in3 and 97% efficiency. Designing a power supply that can blow past the brick

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wall by increasing both density and efficiency is the goal. But it is extremely difficult and only comes with great cost (more expensive components and greater R&D costs).

One truth we have learned is that we have reached the brick wall and it’s not moving. The only way beyond it is to look at circuits differently. So we look at what the circuits are made of and how they behave -- Like using super magnetic materials, or ultra-fast switching semiconductors, or applying switching logic (firmware) to the internal functions of the power supply. Only by looking at circuits from a fresh perspective do we find ways around the brick wall.

Additionally, customers sometimes bring requirements that call for specific operating ranges falling outside what the electronics will support. So the designer may be faced with falling back to an older topology (lower density and lower efficiency) to support the customer criteria.

There is an alternative topology called resonant LLC that takes the bolder approach of looking again at ways in which the electronic circuitry can be manipulated to “have it all.” The goal is to retain maximum efficiency, maximum density and also provide a flexible and resilient power supply that can operate over a wide range of conditions. That is extremely hard to do, yet it is what we have done.

To understand the approach, first consider that one way switch-mode supplies boost efficiency is by becoming smaller. The main way they get smaller is by using higher operating frequencies to reduce the size of the power magnetics and LC components in the output. However, going to higher switching frequencies leads

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ENERGY EFFICIENCY

Current flow through the two inductors during the switching sequence of the two switching transistors Q1 and Q2. The top waveforms are the turnon pulses to the two switching transistors respectively with the resulting voltage at point Vd in the accompanying schematic. The currents through the two inductors ILr and ILm respond as shown, resulting in the current flows though diode bridge legs D1 and D4, and D2 and D3.

to higher transistor switching losses at turn-off and turnon when the supply works with conventional square-wave current. The turn-on losses arise from charging the transistor output capacitances and can be appreciable at frequencies over about 1 MHz. The overlap of falling current and rising collector voltage at turn-off brings a spike of dissipation during turn-off. This effect is more problematic as switching frequencies rise because the more frequent spikes lead to higher average transistor dissipation.

A switching topology called a resonant converter was developed to reduce the amount of energy dissipated in the switching transistor during turn-on and turn-off. Resonant converters associate an LC circuit with the switching transistor to change its current waveform from square to sinusoidal. The switching transistor is timed to turn on and off at the zero crossings of the current sine wave. Thus there is no overlap of falling current and rising voltage at the turnoff, nor rising current and falling voltage at turn-on.

Circuits which turn on and off at zero current are known as zero-current switching (ZCS) types. A point to note, however, is that there can be switching losses at turn-on in a ZCS though there is no overlap of rising voltage and falling current at the zero-crossing of the current sine wave. Zero-voltage switches (ZVS) cope with this difficulty by ensuring the transistor output capacitance is part of a resonant LC circuit. The voltage or energy stored on the capacitor when the transistor is off gets stored as current or energy in the inductor of the resonant circuit. Later in the cycle, the energy gets returned without loss to the power supply bus.

Some switching topologies employ what’s called quasiresonant operation. This refers to switching done in such a way that the transistor turns on when the voltage across it is at a minimum but not zero. The term soft switching is frequently used to refer generally to resonant converter schemes for both ZVS and ZCS as well as to quasi-resonant techniques.

Approaches for soft-switched resonant converters have been around for years, but their realization has sometimes brought mixed results. The problem is it can be tricky to design a circuit able to properly time the turn-on and turnoff of the transistor, typically a field effect transistor. Under different line and load conditions the operating points can shift considerably, and the timing of the turn-on and turn-off must change. But recently, improvements in digital signal processors have enabled more flexible control schemes, making the use of such methods easier.

Typical schematic for an LLC converter. The LLC portion of the circuit is made up by Lr, Lm, and C- . The layout of the LLC is quite close to that of a series-resonant LC converter.

An LC converter might also incorporate a second inductor in the form of a transformer, but the inductance of the second inductor in the LC converter would be set so that the resulting resonance would be far away from the converter operating frequencies. This converter is also shown with a passive diode rectifier bridge in the secondary. Super-efficient LLC converters more typically use MOSFET bridge elements that are actively switched to less power dissipates in the switches themselves.

Not shown in this schematic is the feedback network used to adjust the width of the pulse modulation in response to changes in load.

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Power & Energy Efficiency

One specific ZVS/ZCS approach employs an LLC configuration which uses two inductors in a series-parallel combination with a capacitor.

An ac-dc converter using an LLC topology can reach energy efficiencies approaching 96.5%.

The operation of an LLC converter is often explained in terms of a series-resonant LC circuit combined in series with an additional inductor. The second inductor is in parallel with the load.

The explanation generally begins by first considering the series-resonant circuit by itself. In a general series-resonant circuit, the impedance of the resonant network hits a minimum at the resonant frequency. Because the converter is controlled through frequency modulation, the impedance of the resonant circuit changes with switching frequency which, in turn, changes in response to load changes.

A decrease in load current, for example, tends to decrease the output voltage. A feedback circuit will sense the decrease and move the switching frequency of the converter toward resonance such that more voltage drops across the load. Similarly, a drop in load current results in converter frequency moving away from resonance so more voltage drops across the LC tank circuit.

A point to note about the series-resonant converter is that the current only lags the applied voltage above resonance. An examination of the phase plot for this circuit reveals that below resonance, phase is below 0°, so the network will be capacitive to the input source.

This means the switching transistors won’t operate as zero-voltage switches below this series-resonance point.

One drawback of a series resonant converter is that circuit Q drops as the load decreases. So switching frequency must rise significantly to keep the output regulated. This need for a large frequency change is difficult to implement in practical circuits. In this regard, the LLC resonant converter is designed to overcome the disadvantages of the series resonant converter.

The second inductor in the LLC converter is frequently comprised of the leakage inductance of the primary of a transformer. The load resistance is reflected back into the primary circuit through the turns ratio of the transformer. This action effectively puts the load in parallel with the second inductor.

The leakage inductance value of the transformer is such that it affects the resonance of the LLC components. Specifically, it has the effect of creating a second resonance frequency that differs from that of the series-resonant LC. An examination of the gain and phase plots for the resulting LLC circuit reveals that soft switching is possible only when the LLC circuit operates between the upper and lower resonance frequencies.

Waveforms depicting the drain current Id and drain-source voltage of the MOSFET switching elements show the position in time of zerovoltage switching and zero-current switching.

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ENERGY EFFICIENCY

More specifically, it is possible to show from the operational parameters under various loads that the most advantageous operation point for the LLC converter is at the upper resonance point. At this point, there is little frequency change over an extremely wide range of loads. In fact, the LLC converter can keep the output voltage regulated even when there is no load.

Energy efficiencies in any converter topology can improve through use of an FET synchronous rectifier bridge on the secondary side rather than a diode bridge. Use of an active switching element such as a FET or MOSFET boosts efficiency because these devices have a constant and low resistance when conducting, known as onresistance (RDS(on)). The on-resistance can be 10 mΩ or even lower. The voltage drop across the transistor is then much lower than would be the case for a diode. Thus there is less power dissipated in the switching elements themselves.

All in all, converters using an LLC topology along with synchronous rectification have been shown to work at energy efficiencies of about 96%.

Their ability to operate efficiently across a broad range of power demands and in harsh conditions makes them good candidates for handling industrial applications.

ENERGY. EFFICIENCY. EXCELLENCE. The trusted source for your most critical applications.

Lithium Primary • Lithium Ion • Ni-Cd • Nickel Metal Hydride • Lithium Rechargeable • VRLA • Alkaline

REFERENCES GE Industrial Solutions Business www.geindustrial.com

For more information on how Panasonic can assist with your battery power solution needs, contact us: PHONE: EMAIL: BLOG: WEB:

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Power & Energy Efficiency

DC-DC converters with renewable energy in mind ANKUR PATEL VICOR CORP.

Modern battery, fuel cell and solar cell systems need ways of transferring power in two directions. New dc-dc converters that are bidirectional can eliminate the need for bulky transformers in these applications while bringing more efficient operation.

ALTERNATIVE

PCB prototypes implementing the existing bottom and proposed top BDCs have formats that are dramatically different.

The input voltage is 384 V in the forward direction and 48 V in backward mode. The load current ranges are 0 to 35 A and 0 to 4.5 A in forward and backward directions, respectively.

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energy scenarios involving renewable sources are characterized by the sending of power both up and down utility lines. This bidirectional mode of energy handling has fostered interest in efficient power converters able to handle energy flows equally well in both directions.

In that regard, a new zero-voltage switching (ZVS) bidirectional dc-dc converter (BDC) module has been developed to exhibit higher efficiency and power density, greater reliability and isolation than existing BDCs. At power levels up to 1.65 kW, the 384-to-48-V BDC provides efficiency boosts of 13.5%, 3.4% and 1.5% at loads of 10%, 50% and 100%, respectively.

As a result, the new BDC is a promising choice for fuel cell, solar cell and battery applications. It is also a high-energy density option for hybrid and electric vehicles, UPS, telecom, and power grid applications characterized by a need for high efficiency at both light and at full loads.

Most existing ZVS bidirectional and unidirectional dc-dc converters are low-power devices that can be paralleled to handle high power. Methods to handle the power conversion inefficiencies of such arrangements at light-load or no-load operation fail to deal with the fast and frequent changes in power inputs encountered in renewable power generation, which vary with wind speed and solar intensity. The new BDC will also reduce the number of power conversion steps required in most designs and eliminate the heavy, bulky 50 to 60-Hz power transformer needed to isolate current systems from the ac power grid.

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tlec-mda

DC-DC CONVERTERS

INSIDE A BDC This new BDC is a fixed-ratio dc-dc converter. (Dc-dc converters without output regulation are termed "fixed ratio" because the output voltage is a fixed ratio of the input voltage.) It supports a wide range of input currents and voltages. Its topology is that of a Sine Amplitude Converter (SAC).

Though it has been in use for years, SAC topology may be unfamiliar to some engineers. As a brief review of its operating principles, first consider ordinary dc-dc converters that drive a power inductor or transformer with a square wave. The switch elements (usually MOSFETs) dissipate power during switching, and the square wave is rich with harmonics that must be filtered out.

The output power in this converter is proportional to the converter duty cycle which is controlled via pulse-width modulation to provide more or less power from the transformer secondary. The basic PWM converters can be improved with ZVS and ZCS (zero current switching) by adding a capacitor in the primary circuit to form a quasi-resonant circuit, and by switching the switch elements at the zero-voltage and zero-current crossings.

In conventional PWM ZCS/ZVS schemes, energy in the transformer primary is proportional to 1/2LI2 where L is the primary inductance and I is primary current. Energy transfer is constant for a given circuit configuration. To increase the rate at which energy transfers to the transformer secondary, the switching rate must rise. Thus output

TYPICAL PARALLEL CONFIGURATION

Vin

Vout

A typical approach for handling high-power applications is to parallel several dc-dc converters. But this approach introduces power conversion inefficiencies at light load or no load. Such schemes also don’t work well with the fast and frequent changes in power inputs encountered in renewable power generation applications.

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Line Voltage, Current, and Power – The Basics LINE VOLTAGE Three-phase line voltage consists of three voltage vectors.

• By definition, the system is “balanced” • Vectors are separated by 120° • Vectors are of equal magnitude • Sum of all three voltages = 0 V at Neutral

Line

At any given moment in time, the voltage magnitude is V * sin(α) • V = magnitude of voltage vector • α = angle of rotation, in radians

Poster for free:

ω (rad/s) or freq (Hz)

Line

The resulting time-varying “rotating” voltage vector appears as a sinusoidal waveform with a fixed frequency • 50 Hz in Europe • 60 Hz in US • Either 50 or 60 Hz in Asia • Other frequencies are sometimes used in non-utility supplied power, e.g.

• 400 Hz • 25 Hz

Important to Know • Voltage is stated as “VAC”, but this is really VRMS • Rated Voltage is Line-Neutral • VPEAK = 2 * VAC (or 2 * VRMS ) • 169.7 V in the example below • VPK-PK = 2 * VPEAK • If rectified and filtered • VDC = 2 * VAC = VPEAK AC Single-Phase “Utility” Voltage

120 VAC Example

Volts (Peak), Line-Neutral

200

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At any given moment in time, the current magnitude is I*sin(α) • I = magnitude of current vector • α = angle of rotation, in radians

A

120VAC

150 100 50 0

Phase Angle ω (rad/s) or freq (Hz)

C

THREE-PHASE

Voltage value = VX*sin(α) • VX = magnitude of phase voltage vector • α = angle of rotation, in radians

Voltages can be measured two ways: • Line-Line (L-L) • Also referred to as Phase-Phase • e.g. from VA to VB, or VA-B • Line-Neutral (L-N) • Neutral must be present and accessible • e.g. from VA to Neutral, or VA-N • VL-L conversion to VL-N • Magnitude: VL-N * 3 = VL-L • Phase: VL-N - 30° = VL-L

VA-B

-100 -150 -200

800

Important to Know • Voltage is stated as “VAC”, but this is really VRMS • Rated Three-phase voltage is always Line-Line (VL-L) • Line-Line is A-B (VA-B), B-C (VB-C), and C-A (VC-A) • Line-Line is sometimes referred to as Phase-Phase • VPEAK(L-L) = 2 * VL-L • 679 V in the example to the right • VPK-PK(L-L) = 2 * VPEAK(L-L)

“True” RMS

600 400

0

VRMS =

1 V PK-PK 2 2

VRMS = VAC2 For one power cycle

-800

800

If a neutral wire is present, three-phase voltages may also be measured Line-Neutral • VL-N = VL-L/ 3 • 277 VAC (VRMS) in this example • VPEAK = 2 * VL-N • 392 V in the example to the right • VPK-PK = 2 * VPEAK

Time

AC Three-Phase “Utility” Voltage 480VAC , Measured Line-Neutral

600 400 200 0 -200

B

-400

-800

B

IB

-600

Time

A-N Voltage B-N Voltage C-N Voltage Three-phase Rectified DC

IC

N

IA

C

12

I

9 6 3

Real Power • P, in Watts • = instantaneous V * I for a given power cycle

0 -3 -6

Time A Current B Current C Current

A

Reactive Power • Q, in Volt-Amperes reactive, or VAr • Q = S2 - P2 • Does not “transfer” to load during a power cycle, just “moves around” in the circuit

10 ARMS Example

IA

C

Period 1 Mi = 18 points

B

Period 2 Mi = 18 points

A

mi = point 7

mi = point 25

V B-N

IA

N

V C-N

IB

N V A -N

I

P≠V*I

φ

N

V

Capacitive load

• The digital samples are grouped into measurement cycles (periods) • For a given cycle index i….

• The digitally sampled voltage waveform is represented as having a set of sample points j in cycle index i • For a given cycle index i, there are Mi sample points beginning at mi and continuing through mi + Mi -1.

• Voltage, current, power, etc. values are calculated on each cycle index i from 1 to N cycles.

PTO TA L = VA -N * IA + VB-N * IB + VC-N * IC

IA PTO TA L ≠ VA -N * IA + VB-N * IB + VC-N * IC

V C-N

teledynelecroy.com/motor-drive-analyzer

IRMS

QB

Real Power for each Phase • P, in Watts • = instantaneous V * I for a given power cycle

VRMSi =

IRMSi =

Reactive Power for each Phase • Q, in Volt-Amperes reactive, or VAr • Q = S2 - P2

PB

φ

SB PA

SC

• PTOTAL = PA + PB + PC • STOTAL = SA + SB + SC • QTOTAL = QA + QB + QC

φ φ

QA

SA

PC QC

Line-Line Voltage Sensing Case

B IB VB-C

Current is measured L-N

N

IC

L-L voltages must be transformed to L-N reference:

IA

IB

A

N

IC

VA-N IA

A

VC-A

C

Calculations are straightforward, as described above: • PTOTAL= PA + PB + PC • STOTAL = SA + SB + SC • QTOTAL = QA + QB + QC

B

VB-N

VA-B

C

VC-N

Two Wattmeter Method – 2 Voltages, 2 Currents with Wye (Y or Star) or Delta (∆) Winding

S

Q

φ P

Real Power

Note: Any distortion present on the Line voltage and current waveforms will result in power measurement errors if real power (P) is calculated as |S|*cos(φ). To avoid measurement errors, a digital sampling technique for power calculations should be used, and this technique is also valid for pure sinusoidal waveforms.

Voltage is measured L-L on two phases • Note that the both voltages are measured with reference to C phase

mi + Mi - 1 1 V j2 Mi j=mi

Σ

mi + Mi - 1 1 I j2 Mi j=mi

Σ

Real Power (P, in Watts)

Mathematical assumptions: • Σ(IA + IB + IC) = 0 • Σ(VA-B + VB-C + VC-A) = 0 This is a widely used and valid method for a balanced three-phase system

Pi =

Apparent Power (S, in VA)

Reactive Power (Q, in VAR)

PTOTAL = VA-C * IA + VB-C * IB STOTAL= VRMSA-C * IRMSA + VRMSB-C * IRMSB QTOTAL = STOTAL2 - PTOTAL2

Current is measured on two phases • The two that flow into the C phase

Formulas Used for Per-cycle Digitally Sampled Calculations

VRMS

V A -N

φ

IC

IC

Voltage is measured L-L • Neutral point may not be accessible, or • L-L voltage sensing may be preferred

Inductive load

-9

Delta (∆) 3-phase Connection • Neutral is not present in the winding (in most cases)

C

Apparent Power • |S|, in Volt-Amperes, or VA • = VRMS * IRMS for a given power cycle

-12 -15

IB

Digital Sampling Technique for Power Calculations�

A

V

φ

Single-phase Real, Apparent and Reactive Power AC Three-Phase "Line" Currents 15

A IC

480 VAC Example

N

P≠V*I N

• For inductive loads • The current vector “lags” the voltage vector angle φ • Purely inductive load has angle φ = 90° • Capacitive Loads • The current vector “leads” the voltage vector by angle φ • Purely capacitive load has angle φ = 90°

Important to Know • Current is stated as “lAC”, but this is really IRMS • Line currents can represent either current through a coil, or current into a terminal (see image below) depending on the three-phase winding connection • IPEAK = 2 * IRMS • 14.14A for a 10 ARMS current in the example to the right • IPK-PK = 2 * IPEAK

A-B Voltage B-C Voltage C-A Voltage

Three-Phase Winding Connections

C

Single-phase, Non-resistive Loads

Line Current Measurements

-600

As with the single-phase case, Power is not the simple multiplication of voltage and current magnitudes, and subsequent summation for all three phases.

IB

Apparent Power for each Phase • |S|, in Volt-Amperes, or VA • = VRMS * IRMS for a given power cycle

For capacitive and inductive loads • P ≠ V * I since voltage and current are not in phase

-400

B

For one power cycle

A

120°

Current value = IX*sin(α) • IX = magnitude of line current vector • α = angle of rotation, in radians

200

-200

Line-Neutral Voltage Measurements

Wye (Y) 3-phase Connection • Neutral is present in the winding • But often is not accessible • Most common configuration

VPK-PK

Neutral

Like voltage, the resulting time-varying “rotating” current vectors appear as three sinusoidal waveforms • Separated by 120° • Of equal peak amplitude for a balanced load

AC Three-Phase “Utility” Voltage 480VAC , Measured Line-Line

480 VAC Example

Time

“Not True” RMS

Power Factor (PF, or λ) • cos(φ) for purely sinusoidal waveforms • Unitless, 0 to 1, • 1 = V and I in phase, purely resistive load • 0 = 90° out of phase, purely capacitive or purely inductive load • Not typically “signed” – current either leads (capacitive load) or lags (inductive load) the voltage

C

VC

Three-phase, Non-resistive Loads

For purely resistive loads • PA = VA-N * IA • PB = VB-N * IB • PC = VC-N * IC • PTOTAL = PA + PB + PC V B-N

Voltage Current

120°

120°

VA-N

Line-Line Voltage Measurements

V

Resistive load

Three-phase, Resistive Loads

Three-phase, Any Load

ω (rad/s) or freq (Hz)

Like voltage, three-phase current has three different line current vectors that rotate at a given frequency • Typically, 50 or 60 Hz for utility-supplied voltage

VA

N

B

By definition, the system is “balanced” • Vectors are separated by 120˚ • Vectors are of equal magnitude • Sum of all three currents = O A at neutral (provided there is no leakage of current to ground)

Neutral

120°

P=V * I

I

Power Factor

Phase Angle (φ) • Indicates the angular difference between the current and voltage vectors • Degrees: - 90° to +90° • Or radians: -π/2 to + π/2

Line

Neutral

The resulting time-varying “rotating” voltage vectors appear as three sinusoidal waveforms • Separated by 120° • Of equal peak amplitude

If all three phases are rectified and filtered • VDC = 2 * VL-N * 3 = VPEAK * 3 = 679 V in the example to the right

-50

For purely resistive loads • P = I2R = V2/R = V * I • The current vector and voltage vector are in perfect phase

120°

VB

Neutral

to receive a Power Basics

Neutral

120°

The three voltage vectors rotate at a given frequency • Typically, 50 or 60 Hz for utility-supplied voltage The single-phase voltage vector rotates at a given frequency • Typically, 50 or 60 Hz for utility-supplied voltage

120°

ω (rad/s) or freq (Hz)

Typically, the three phases are referred to as A, B, and C, but other conventions are also used: • 1, 2, and 3 • L1, L2, and L3 • R, S, and T

THREE-PHASE

Electric Power • “The rate at which energy is transferred to a circuit” • Units = Watts (one Joule/second)

The resulting time-varying “rotating” current vector appears as a sinusoidal waveform

Imaginary Power

Neutral

MDA800 Series Motor Drive Analyzers 8 channels, 12-bits, 1 GHz

SINGLE-PHASE

Like voltage, the single-phase current vector rotates at a given frequency • Typically, 50 or 60 Hz

Line Current (Peak)

• Magnitude (voltage) • Angle (phase) Typically, the single-phase is referred to as “Line” voltage, and is referenced to neutral.

LINE POWER

SINGLE-PHASE B

Volts (Peak), Line-Line

MDA800 Series and sign up

THREE-PHASE

Single-phase line voltage consists of one voltage vector with:

Volts (Peak), Line-Neutral

Learn more about the

LINE CURRENT

SINGLE-PHASE

mi + Mi - 1 1 Vj * Ij Mi j=mi

Σ

B

B

IB VB-C

C

Power Factor (λ)

N

IA VA-C

A

IB

VB-C

C

λi =

VA-C

A

IA

Pi Si

Si = VRMSi * IRMSi

magnitude Qi =

S i2 - P i2

Sign of Qi is positive if the fundamental voltage vector leads the fundamental current vector

Phase Angle (φ)

magnitude Φi = cos-1λi Sign of Φi is positive if the fundamental voltage vector leads the fundamental current vector

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Power & Energy Efficiency

power depends directly on switching frequency.

Switching frequencies in ZCS/ZVS resonant converter have a trade-off between stored energy per cycle and the conditions needed to realize ZVS/ZCS. SAC topology resembles that of ZCS/ZVS but works differently. SACs operate at a fixed frequency equal to the resonant frequency of the primary side tank circuit -- there is no PWM. The frequency is fixed regardless of load on the secondary.

When the load increases, the SAC reacts by boosting the amplitude of the sinusoidal current on the primary resonant tank, increasing the energy coupled into the secondary. The FETs in the SAC primary circuit switch at zero-crossing points, and the current in the primary resonant tank is a pure sinusoid, greatly reducing harmonic content. Unlike ordinary ZCS/ZVS circuits, the transformer leakage inductance in SAC is not a critical energy storage element. So the SAC can operate at much higher frequencies, allowing the use of a small transformer.

To understand how an SAC operates, it is useful to describe its primary circuit in more detail. The SAC creates a low-voltage sinusoidal oscillation by resonating the small amount of leakage inductance present in the power transformer with a primary-side capacitor. The amplitude of this oscillation varies with the current

drawn by the load. As the load current rises, the amplitude of the oscillation (as measured across the resonant capacitor) also rises. When the load draws no current, the amplitude drops to zero.

The SAC topology is also useful because it is inherently bidirectional. The SAC topology incorporates synchronous rectification on the secondary side of the transformer, so the converter can process power “backwards,” from the output back to the input. In other words, it can both source current to the load and sink current away from the load. As a result, whenever the output voltage exceeds the input voltage multiplied by the transformer turns ratio, power will flow from the output to the input. Reverse power flow is efficient: Energy taken from the output is delivered to the SAC input, except for incidental losses, such as the loss in parasitic circuit resistances.

ANALYZING OPERATIONS In the case of the BCS, the primary circuit is a half-bridge that is stacked (high side and low side devices mounted on opposite sides of a common conductive interface).

The primary also uses low-voltage MOSFETs to reduce conduction losses and to realize low switching losses across the entire load range because of ZVS/ZCS technology.

The secondary circuit is center-tapped with synchronous rectification. A high-frequency transformer provides galvanic isolation. In ideal conditions, its voltage and current ratios are defined by the following equation, with K being the transformer’s turns ratio.

VOUT

Tank circuit

VIN

SIMPLE SAC Primary

=

IOUT IIN

=K

Secondary

When the load increases, the SAC reacts by boosting the amplitude of the sinusoidal current on the primary resonant tank, increasing the energy coupled into the secondary. The FETs in the SAC primary circuit switch at zero-crossing points, but the current in the primary resonant tank is a pure sinusoid.

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A53_ED_FullPg_9x10_875_Layout 1 8/26/16 10:31 AM Page 1

DC-3 Series

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High Input Voltages Up to 900 VDC • DC-1 Series

• 120-370 VDC input voltage range • 5-300 VDC regulated isolated outputs • Up to 300 watts output power

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• 300-900 VDC input voltage range • 3.3-300 VDC regulated isolated outputs • Up to 50 watts single & dual outputs

• HiQP Series

• 125-475 VDC input voltage range • 24-200 VDC regulated isolated outputs • Up to 50 watts output power

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Power & Energy Efficiency

The BDC employs a 1.1 MHz switching frequency.

The input capacitors (CIN) are specified to be identical so they balance the voltage in the stacked half-bridge primary circuit, which facilitates storing the same energy in each primary winding. The output voltage is proportional to the input voltage minus the voltage drop due to the combined resistance of the MOSFET’s RDS-ON , the PCB trace and transformer’s winding resistance.

The new BDC uses a planar transformer for low losses and galvanic isolation. The bidirectional BDC provides step-down dc-dc conversion when operating from input-to-output terminals – forward mode. It’s implemented by connecting a highvoltage source to the input terminals and a lowvoltage load to the output terminals.

To enable backward mode, the BDC first must be turned on in forward mode, because SAC control is on the input side. This turn-on takes place by applying a low voltage to the input terminals.

SIMULATION: TYPICAL PARALLEL APPROACH

Then, when the voltage applied to output terminals exceeds the turns ratio times the input voltage, the BDC enters backward mode, providing step-up, lowvoltage-to-high-voltage dc-dc conversion from the output terminals to the input terminals.

A start-up circuit based on flyback topology, receiving its input from the low-voltage side, may be used to enable backward power flow. Then, the high voltage side provides the bias voltage and current to SAC control. The low line of the BDC on the high voltage side is 260 V. There is an under-voltage turnon point below the low line where the BDC turns on.

The new BDC, with a transformation factor (K) of 1/8, is used to implement 384-to-48-V bidirectional conversions.

Two PCB prototypes have been designed implementing the BDCs. The input voltage is 384 V in the forward direction and 48 V in backward mode.

The load current ranges are 0-to-35 A and 0-to-4.5 A in forward and backward directions, respectively.

SIMULATION PERFORMANCE Parameters

Existing BDC

New BDC

Efficiency (%)

94.77

97.94

Power loss (W)

45.88

17.49

Footprint area (cm2)

42.90

14.44

Solution cost ($)

339.78

247.04

SIMULATION: BDC

Simulations conducted with Vicor’s power-bench whiteboard tool show how existing and new BDCs behave. Efficiency and power loss are analyzed for both BDCs at 384 Vdc input, 50% load and 25°C operating temperatures

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DC-DC CONVERTERS

In one case where the new BDC is used, a dc micro-grid has a 384-V high-voltage dc (HVDC) side and low-voltage dc (LVDC) side at 48 V, 24 V and 12 V. The external 230 Vac grid is rectified and converted to 384 Vdc for the HVDC bus by one centralized PFC boost rectifier. The BDCs will convert HVDC to the lower dc voltages. The centralization improves efficiency by eliminating redundant conversions.

A BOON FOR ALTERNATIVE ENERGY The new BDC can interface the low-voltage generated by solar cells to the HVDC bus. The bidirectional PFC boost rectifier can shift excess power generated by the home back to the power grid. Excess power can also be stored locally by using the proposed BDC to lower the HVDC to storage system levels. Similarly, the high efficiency, compact size, light weight and high reliability of the new BDC are vital for hybrid vehicles.

The same can be said for ac/dc grid systems. The BDC eliminates the need for a heavy, bulky 50 to 60 Hz power transformer by effecting galvanic isolation and voltage matching at 1.1 MHz. For bidirectional power processing in energy storage systems, the proposed BDC would provide current from a 380-V bus to charge a battery bank in the forward direction or to provide energy from the battery bank backward to the high-voltage bus. In one case, two post regulator modules (PRMs) regulating against line and load and are implemented using Vicor ZVS buck-boost regulator modules.

PERFORMANCE GAINS OF NEW BDC Parameters

Existing BDC

New BDC

Number of converters

6

1

Input Voltage Range (V)

360-400

260-410

Efficiency at 10% load (%)

81

94.5

13.5% better

Efficiency at 50% load (%)

94.4

97.8

3.4% bettter

Efficiency at 100% load (%)

95.7

97.2

1.5% better

Output resistance (mΩ)

28.3

22.6

Lower resistive losses

No load power dissipation (W)

39

10

Lower no load losses

Volume (cm )

28.86

10.48

Occupy less PCB space

Power density (W/cm3)

57.2

157.4

2.75 times more

Weight (g)

84

41

Weight is half

3

Gains of New BDC Less number of converters Wider input voltage range

A summary of how traditional and new BDCs perform.

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REFERENCES

VI Chip BCM Bus Converter Module Data Sheet, [online] Rev1.9, 04/2011 www.cdn.vicorpower.com/documents/datasheets/ BCM384_480_325A00.pdf Maxi dc-dc Converter Data Sheet [online] Rev9.2, 09/2014 www.cdn.vicorpower.com/documents/datasheets/ds_375vinmaxi-family.pdf Chip BCM Bus Converter Module Data Sheet [online] Rev1.3, 05/2015 www.vicorpower.com/documents/datasheets/ds_ BCM400P500T1K8A31.pdf VIA BCM High Voltage Bus Converter Module Data Sheet [online] Rev1.1 06/2015 www.vicorpower.com/documents/datasheets/ds_ BCM4914xD1E5135yzz.pdf “Vicor Factorized Power Architecture and VI Chips,” Vicor White Paper www.vicorpower.com/documents/whitepapers/fpa101.pdf www.us.tdk-lambda.com/ftp/brochures/TDK_TJ023_DCDC_E_1105.pdf Webb, Victor-Juan, "Design of a 380 V/24 VDC microgrid for residential DC distribution" (2013). Theses and Dissertations. Paper 231.

Sonya Gargies, Hongjie Wu and Chris Mi, “Isolated Bidirectional dc-dc Converter for Hybrid Electric Vehicle Application” (2006) Fan Haifeng, “Advanced Medium-Voltage Bidirectional dcdc Conversion Systems for Future Electric Energy Delivery and Management Systems” (2011) Electronic Theses, Treatises and Dissertations. Page 4507.

Hamid R. Karshenas, Hamid Daneshpajooh, Alireza Safaee, Praveen Jain and Alireza Bakhshai, “Bidirectional dc-dc Converters for Energy Storage Systems, Energy Storage in the Emerging Era of Smart Grids, Prof. Rosario Carbone (Ed.), ISBN: 978-953-307-269-2, InTech, Available from: www.intechopen.com/books/energy-storage-in-the-emergingera-of-smart-grids/bidirectional-dc-dcconverters-for-energystorage-systems Xiaoyan Yu and Paul Yeaman, “A new high efficiency isolated bi-directional dc-dc converter for DC-bus and battery bank interface,” in 2014 Applied Power Electronics Conference VI Chip PRM Pre Regulator Module Data Sheet, [online] Rev1.2, 08/2013 www.cdn.vicorpower.com/documents/datasheets/ PRM48BF480T500A00_ds.pdf

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Power & Energy Efficiency

Bumpless control for power factor correction It can be difficult to correct for power factor in converters that have different conduction modes.

A special technique can make corrections smoothly.

JOEL STEENIS, ALEX DUMAIS MICROCHIP TECHNOLOGY, INC.

DESIGNERS

of power factor correction (PFC) circuits will invariably notice distortion as the input current approaches zero. The distortion problem becomes progressively worse at light loads. It arises because of the two fundamental operating modes of a power converter and because the current measurement signals are below the noise floor. The two operating modes of a converter are continuous conduction mode (CCM) and discontinuous conduction mode (DCM). The distinction between the two operating modes is made based on inductor current. In CCM the inductor current is a triangle-shaped waveform. In DCM the inductor current waveform is triangular shaped but has “flat spots” at zero current.

A technique called bumpless control addresses the issue of alternating between CCM and DCM, is practical to implement, and reduces total harmonic distortion (THD) in hardware.

Input Voltage

INDUCTER CURRENT IN CCM AND DCM

Time A hypothetical discontinuity “bump” may present itself this way in the output of a switched controller that transitions between DCM and CCM operation.

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BUMPLESS CONTROL

The boost converter is a common building block for a number of PFC topologies and has different small-signal characteristics for each operating mode. Small-signal models GiCCM(S) and GiDCM(S) for CCM and DCM respectively in each operating mode are:

(1)

Gi CCM (S) =

(2)

Gi DCM (S) =

îL d

îL d

=

RC +1 2 LC L S2 D' 2 +S RD' 2 S

2V OUT RD'

2

4MV =

IN

RLC S+

2M-1 (M-1)RC

S

+1

RC +1 2 S+

2(M-1) D1TS

TYPICAL CCM, DCM BODE PLOT

Magnitude (dB)

where R is the load resistance, C is the output capacitance, L is the boost inductance, M is the voltage conversion ratio (VOUT/VIN), Ts is the switching period, D1 is the duty cycle of the converter in DCM, D’ is the complement of the duty cycle (D’ = 1-D) in CCM, VOUT is the output voltage, VIN is the input voltage, î L is the small-signal variation in inductor current, d is the small signal variation in duty cycle, and s is the frequency in rad/sec.

An examination of the Bode plot for a representative system makes it clear that a single controller will yield quite different loop gains and result in different responses for each operating mode. A qualitative comparison of the two plant qualities shows that a controller designed for a PFC operating in CCM will result in a low-bandwidth loop when operating in DCM. To address the CCM/DCM issue, one may instinctively choose to use two controllers. In principle, this approach is valid; however, it will invariably lead to a discontinuity when switching between controllers. To resolve the discrepancy in controller outputs, i.e. controller “bump,” the controller structure must be modified.

One approach is to add a sub-controller to each controller in the original configuration. Also added are complementary switches which control each subcontroller, making it active or inactive. The switches are additionally configured to force the output of the inactive controller to track the output of the active controller.

Readers will note a potential source of confusion: There are controllers in the controller. The concept

of “the controller” becomes ambiguous and notation is of utmost importance. For clarity, we introduce the term “master controller.” This is the controller with input and output always connected to the system. The “master controller” may contain any number of subcontrollers and switches, connected in a variety of ways.

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Frequency (Hz) A typical Bode magnitude plot for a PFC in CCM and in DCM regions.

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Power & Energy Efficiency

In the case of a DCM controller, the signal “eDCM bump” is the difference between the output of the DCM controller and CCM controller. The bump removal controller is active for the DCM controller. (It should be forcing the output of the DCM controller to track the output of the CCM Consider two cases that refer to the accompanying figures. controller.)

The DCM controller does not CASE 1: SWITCH SDCM IS CLOSED AND SCCM IS OPEN operate normally. The output of In the case of a DCM controller, the signal “eDCM bump” is zero. The the CCM controller is now the input bump removal loop is inactive for the DCM controller, and the DCM for the DCM controller and the controller operates normally. input to the bumpless controller In the case of a CCM controller, the signal “eCCM bump” is acts as a disturbance. the difference between the output of the CCM controller and DCM To summarize, there are four controller. The bump removal controller is active for the CCM controller.

(It should be forcing the output of the CCM controller to track the output feedback loops to be considered in the bumpless controller scheme: of the DCM controller.) the plant/DCM controller loop, The CCM controller does not operate normally. The output of the DCM controller is now the input for the CCM controller, and the input to plant/CCM controller loop, DCM controller/KBUMP DCM loop, and the the bumpless controller acts as a disturbance.

CCM controller/KBUMP CCM loop.

CASE 2: SWITCH SCCM IS CLOSED AND SDCM IS OPEN The controllers for each of these loops may be synthesized using In the case of a CCM controller, the signal “eCCM bump” is zero. The pole placement, K-factor, or other bump removal loop is inactive for the CCM controller, so the CCM methods. The design objectives controller operates normally.

Simply put, the bumpless controller switches between subcontrollers, each with their own feedback loop. The sub-controllers are designed to track either the input to the bumpless controller or the output of the bumpless controller depending on the position of two switches, SCCM and SDCM. The switches control which sub-controller is active or inactive and force the output of the inactive controller to track the output of the active controller.

Switched Controller (Master Controller)

Output of Switched Controller

One way to realize a converter operating in DCM and CCM is with a controller that switches between two “sub-controllers,” one for each operating mode. The problem: This switched controller (i.e.

“Master Controller”) does not ensure continuity when switching between the DCM and CCM controllers and may have a “discontinuity bump” at switching instants.

A hypothetical discontinuity “bump” may present itself this way in the output of a switched controller that transitions between DCM and CCM operation.

Time at which DCM/CCM transition occurs

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BUMPLESS CONTROL

for the plant/DCM controller and plant/CCM controller loops are germane and addressed in many references.

Conversely, design objectives for the DCM controller/KBUMP DCM loop and the CCM controller/KBUMP CCM loop are opaque. Given the DCM and CCM controllers (sub-controller 1 and sub-controller 2), one must consider the output of either controller near the time of transition. Given the frequency and shape of the signal to be tracked, one may define a structure (PID, two pole/two zero, etc.) for KBUMP CCM and KBUMP DCM that will result in optimal performance for the given resources.

To see bumpless control in an actual application, consider the example of a 750-W semi-bridgeless PFC. The PFC is based on the Microchip dsPIC33EP64GS502 microcontroller and fits into a standard 1U form factor. It was designed to meet the CSCI (Climate Savers Computing Initiative) titanium efficiency spec.

Both the CCM and DCM controllers are type 2 and use constant-gain bump removal controllers. Computational complexity is always a concern when implementing advanced control algorithms. With the current control loops operating at 100 kHz, the bumpless controllers use 30 cycles more (4.3% more MIPS) than a single CCM controller.

One will note from the accompanying Bode plots for this example that the loop gain in DCM is lower than the loop gain in CCM. This is to guard against instability in the case of CCM/DCM detection issues, while still providing greater bandwidth then can be had from a CCM controller when the plant is in DCM. The PFC was simulated using the SimPowerSystems toolbox in Matlab Simulink. The simulation predicted a THD of 16.5% at light loads when using a CCM controller and a THD

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Power & Energy Efficiency

BUMPLESS CONTROLLER

of 8.9% at light loads when using a bumpless controller.

A review of performance specifications reveals that the bumpless controller reduces THD by approximately 2% at light loads. This is a marked improvement, given that little effort was devoted to tuning the controllers for optimal performance, and the CCM/DCM detection scheme detected the transition earlier than expected.

All in all, the absence of bumpless control leads to PFC circuits that have distortion as a result of two distinctly different operating modes. Simply switching between two controllers addresses the plant response in each mode but not the transition between the two modes. Adding the bumpless controller addresses the plant response in each operating mode as well as the transition. Bumpless controllers are practical to implement and reduce PFC THD in hardware.

A bumpless controller incorporates four “sub-controllers.”

The switches SCCM and SDCM are complementary and denote which sub-controller is active or inactive. The switches are also configured to force the output of the inactive controller to track the output of the active controller.

REFERENCES Microchip Technology Inc. www.microchip.com

CONTROLLER STATUS CASE 1

CCM Controller

eCCM bump

KBUMP CCM

SCCM

Controller

Percent Load

THD (%)

Active

Zero

Inactive

On

CCM Controller

10

12.2

DCM Controller

eDCM bump

KBUMP DCM

SDCM

-

15

14.2

-

20

8.1

Inactive

Non-Zero

Active

Off

-

100

2.2

Bumpless Controller

10

10.8

-

15

12.2

-

20

6.4

-

100

2.2

CASE 2

CCM Controller

eCCM bump

KBUMP CCM

SCCM

Inactive

None-zero

Active

Off

DCM Controller

eDCM bump

KBUMP DCM

SDCM

Active

Zero

Inactive

On

Here’s a comparison of PFC THD using a single CCM controller and a bumpless controller.

Status of bumpless controller parameters when switches SCCM and SDCM are oriented for CCM and DCM controllers.

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BETTER THERMAL DESIGN

Better thermal design means better efficiency The strategic use of computational fluid dynamics can speed the development of high-tech products.

BORIS MAROVIC, JOHN PARRY

MENTOR GRAPHICS CORP.

TRADITIONALLY,

analysis using computational fluid dynamics (CFD) took place late in the design process once changes to part geometries had pretty much stopped. But CFD is starting to be applied at earlier stages of development.

For an indication of the trend, consider recent events at Blue Origin, the spaceflight services company set up by Amazon.com founder Jeff Bezos. The firm’s BE-4 rocket engine, scheduled to power Blue Origin’s rockets by 2019, has undergone several million hours of CFD modeling to perfect preburner sizing and injector element designs.

Bezos credits CFD with significantly shortening the engine’s design and testing schedule.

Of course, it should be noted that CFD is not something used just by rocket scientists.

The latest tools let design engineers do CFD analysis. This means routine thermal simulation and analysis can take place earlier in the design cycle. The early application helps to reduce design reiterations, improve quality, and shorten the time to get products to market.

Here are a few steps that electronics companies can use to boost productivity when applying CFD while also making it easier for engineers to design complex systems and components.

Let designers do routine CFD -- In a workflow where designers routinely handoff design baselines to a CFD analysis department, it can take days, sometimes weeks, for results to come back. Meanwhile, the design will have evolved and, if there are fundamental changes necessary to address thermal issues, a lot of work will have been wasted. However, in many cases, designers can undertake routine CFD to check the effect each design iteration has on product performance without having to send it out of their department.

Let CFD analysts do the most complex CFD work -- CFD specialists tend to use one high-end CFD tool for all simulations, regardless of how simple or complex

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Headlight modeled in FloEFD CFD software, illustrating the meshing underneath the image.

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Power & Energy Efficiency 38

the job. However, 80% of simulations don’t need an analyst’s in-depth expertise. Tying up analysts with routine work limits the usefulness of their highly valued experience. Also, the specialized tools they use are expensive and in short supply.

Specialists tied up with pre- and post-processing tasks or dealing with mundane projects can’t help designers.

Keep up-to-date on work practices -- Despite enormous advances in tools and technology, people are surprisingly slow to change their own work habits, including how departments are organized. Mature companies often have design and analysis departments set up years ago to manage a historical workflow in a way that is no longer optimum. With today’s market demands, these old structures can impede progress. Before a design engineer can hand off thermal analysis to the CFD specialist, he or she must create a formal specification of the analysis task, including the detail expected and reporting format of the results.

Even as technology has evolved, many designers still believe CFD tools are too difficult and time-consuming to learn and use. Many managements and/or in-house analysts also still believe designers can’t do a reasonably accurate analysis in a timely manner. This idea is outdated and wrong.

Modern tools give designers fast “directional” guidance about how to improve designs, and many smaller, more flexible, companies are embracing them. As well, some larger companies are designing new workflows using modern tools because they must reduce costs when developing new products.

Interact more, report less -- One of the most time-intensive activities when doing CFD with traditional tools is the meshing. It can account for as much as 70% of the time spent on CFD. For example, in one case the meshing of an LED headlight spanned about 2.5 weeks, whereas a tool such as FloEFD can handle the task in 15 minutes. The solver time is then around 10 to 20% of the process -- anywhere from hours to a few days -- depending on the item simulated, the physics considered, and whether conditions are stationary or transient.

It also takes time for specialists to understand and interpret simulation results in the context of the desired product performance, then feed results and recommendations back to designers.

Formal reporting bloats an already time-intensive process. The reporting process must be balanced against the value of the simulation results and the

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possibility of the design baseline used for the simulation becoming stale with lengthy delays.

As products become more complex, it’s harder to make design trade-offs between performance goals and considerations such as weight and cost. It’s crucial that engineering interact with other departments because real-time communication speeds the workflow. When formal reporting is needed, use tools that automate reporting from data such as Microsoft Excel and Word files, with customizable templates to save time.

Use CFD more often earlier in the design – In many companies CFD has become an activity that only happens late in design, with one detailed CFD study before the product goes into production or to physical prototyping. However, conducting routine CFD within the design workflow improves the quality of the product, often resulting in fewer or no physical prototypes. And it forces the CAD geometry for the final CFD analysis to be prepared well in advance.

Create consistent quality -- CFD analysts craft meshes and can adjust many little-known functions in their traditional body-fitted CFD tools. So if the software has 20 different turbulence models, it can produce at least 20 different results, all other things being equal. In comparison, modern CFD tools that use Cartesian-based meshes are not only simpler but, because they intelligently provide guidance through the analysis workflow, give results with far less unnecessary variability, again saving time and resources.

Get quick simulation results -- Timeliness is crucial in making good decisions. Thermal analysis should focus on getting good answers. A result that is sufficiently accurate and timely enough to support a proposed design change is more useful than a better answer delivered too late.

CFD tools should be able to turn around a simulation within 24 hours. That turn-around time should cover the use of modified CAD geometry, performing meshing, creating a solution, and postprocessing the results to avoid slowing the overall design workflow.

Improve overall simulation accuracy -- CFD traditionally is time-consuming, particularly for CAD preparation and meshing stages. It often takes too long to create a good-quality fine mesh, so engineers instead use coarse low-quality meshes for simulation. This practice results in high levels of mesh-generated error that degrade the quality of the simulation results.

However, new techniques let CAD geometry mesh quickly, reliably, and without over-simplification. These newer solutions are augmented by technologies that make simulations on coarse meshes more accurate by using, for example, empirical approaches to predict pressure drop and heat transfer in narrow channels that are not adequately meshed.

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THERMAL DESIGN CHALLENGES

Liberate the design space -- The earlier CFD can be used in design, the greater the chance engineers can confirm the product’s performance and make it even better.

Exploring the product’s potential around a single design point almost always leads to the discovery of a better design or to improvements to the current design. These investigations can also reveal which aspects of the design are crucial to product performance.

Eliminate CAD data conflicts -- When adopting new CFD tools, pay special attention to the CAD data format they can use. It’s best if the thermal simulation and analysis tool can use native CAD data; it is far easier to fix issues than when working with neutral file formats such as IGES and STEP. When non-native data is imported, it is as detailed as native data, but often the import results in interpretation errors. The problem manifests itself as missing surfaces and holes in the models that must be healed or fixed. This model mending takes time and sometimes is not possible at all if too much of the geometry is lost during import.

Traditional body-fitted CFD tools often use geometry healing tools or a surface wrapper to (hopefully) fix such imported geometries. But it is often a lot of work to produce a healthy geometry for the CFD simulation. Complex geometries are usually simplified to make meshing easier, and this simplification can degrade the realism of the simulation.

CAD data that must be simplified or re-imported through neutral file formats also introduces other problems. For example, it’s tough to investigate the use of parts that are outside general parameters or how changes to parts can affect the design when using CAD data in neutral file format.

Working with designers who can use CFD in their process, we noticed a faster feedback loop with naturally built native CAD data, and better CFD results.

The future is interdependent -- Today’s CFD software, such as Mentor Graphics FloTHERM XT and FloEFD, lets thermal, mechanical, and electrical engineers work more effectively with CFD analysts to create better products quickly.

Design engineers can “front load” analysis, determine trends, accurately analyze and solve problems faster and earlier in the design process. They can complement what specialists must do in the later stages of verification. This close interaction and involvement can reduce thermal simulation and analysis time from weeks or days to hours.

CFD can let design engineers test various options using a design-of-experiments approach to arrive at a more competitive and/or reliable product. Both designers and CFD experts can use modern fast and accurate thermal simulation and analysis tools to more efficiently work together creating the complex electronics that modern products demand.

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REFERENCES Mentor Graphics Corp. www.mentor.com

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Power & Energy Efficiency

Customized approach to power efficiency VPX power systems are a defense industry mainstay. New standards help boost efficiency

GERALD HOVDESTAD BEHLMAN ELECTRONICS

by eliminating the need to over-specify supplies when handling outlier situations.

POWER

Supplies have always been the problem child in electronic system design. Often, high-end systems are configured for specialized applications. As a result, power requirements tend to be quite different from one configuration to another. VITA (formerly the VMEbus International Trade Association) attempted to standardize some aspects of VME system design, which also put constraints on power. Mechanical standards were agreed upon for circuit cards, and there was an effort to organize voltages and such qualities as

regulation, noise, ripple, and rise time. There was really no successful attempt to control the size and shape of VME power supplies.

After the introduction of the VPX standard VITA 46, the standardization process continued to advance and define many more features of system cards. Chassis, backplane, and card profiles are defined in VITA 65, although this standard appears to be a monster with unlimited growth potential.

In addition, the VITA 62 specification was written in an effort to standardize power supplies. This specification defines mechanical configurations, connectors, voltages and general interfaces, so system integrators can design platforms with standard backplanes and can plug in power supply cards from a variety of suppliers.

The expectation was that cards from these different manufacturers would function properly and identically. This expectation may be true with some less complex systems, but there are still many variables, such as user-defined pins,

VME power supplies have historically come in many different sizes and shapes, as is evident in this selection of supplies.

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CUSTOMIZED APPROACH

which allow designers to make use of sophisticated, “undefined” features.

These features are specification enhancements offered by some power supply manufacturers but not by others, even if they are VITA 62 compliant.

A further complication is the programmability often offered with these power supplies, which is loosely defined in the VITA 46.11 system management specification.

VITA 62 does define a starting point that system designers can use for basic power requirements. They must add actual power levels, as well as decide how to incorporate other undefined system power needs.

EMI filtering and hold-up requirements, as well as transient and lightning protection, are just some of the possible considerations. If multiple cards are used, issues that must be resolved include shared circuit performance, compatibility, sensing, and synchronization.

The VITA 62 specification, as currently released, allows for considerable flexibility in requirements. Output currents for each allowable voltage are the parameters that are most variable. Seldom does a power supply company see two requirements with the same output power levels. So it is important that products can be tailored to meet these needs. System integrators should not be afraid to ask for what is important, and power supply manufacturers must be able to discuss these needs and offer products that will meet them. Sometimes it might be important to stick with a fully compliant power supply, while other programs may allow some configuration changes in the interest of space savings or cost.

An example is a system with a 5-V, 120-A requirement but which needs less power on the 12 and 3.3-V outputs. This type of situation is fairly common because some processor assemblies, widely used in VPX systems, need a significant amount of 5-V power. Standard 3U VPX cards come with outputs up to 5 V at 40 A because the pins designated for the 5-V output are rated at 40 A. One system designer may decide to meet the requirement using three cards and stay fully compliant with the VITA 62 specification. Another may opt for some tailoring of the card to reduce size and cost. The latter system designer may settle on using one standard card and a second card with two 5-V outputs by replacing the 12 V on that card with a second 5-V model.

This approach would save a card slot, and it would cut the cost of the power supply roughly 33% because it used fewer cards.

Each program designer must decide what degree of specification compliance or deviation they’ll accept. The designer should not be reluctant to discuss the requirement with the supplier; this interaction is usually the best way to optimize the design. In fact, it is important to consult with the power supply manufacturer early in the design stage because a reconfiguration of the power supply could affect other parts of the system design. The result could be added cost for reworking and more development time.

Besides hardware requirements, VPX power supply software capability is also important for customizing and optimizing systems. Modern power supplies can be quite flexible. While IPMI-based VITA 46.11 can be used to optimize system performance, it doesn’t do much for defining programmable functions. Some manufacturers continue to use a PMBus-based I2C structure because the functions available are well defined.

The specification for three-phase inputs on 3U cards. The current 3U connector does not have enough input pins to handle basic logic and control functions such as “card ready” or “fault.”

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Pin Number

Rated Current (A)

Pin Name

P1

40A

-DC_IN/ACN

P2

40A

+DC_IN/ACL

LP1

20A

CHASSIS

A1

<1A

UD1

B1

<1A

UD2

C1

<1A

UD3

D1

<1A

UD4

A2

<1A

VBAT

B2

<1A

FAIL*

C2

<1A

INHIBIT*

D2

<1A

ENABLE*

A3

<1A

UD0

B3

<1.5A

+12V_AUX

C3

<1A

NED

D3

<1A

NED_RETURN

A4

<1.5A

3.3V_AUX

B4

<1.5A

3.3V_AUX

C4

<1.5A

3.3V_AUX

D4

<1.5A

3.3V_AUX

A5

<1A

GA0*

B5

<1A

GA1*

C5

<1A

SM0

D5

<1A

SM1

A6

<1A

SM2

B6

<1A

SM3

C6

<1.5A

-12V_AUX

D6

<1A

SYSRESET*

A7

<1A

PO1_SHARE

B7

<1A

PO2_SHARE

C7

<1A

PO3_SHARE

D7

<1A

SIGNAL_RETURN

A8

<1A

PO1_SENSE

B8

<1A

PO2_SENSE

C8

<1A

PO3_SENSE

D8

<1A

SENSE_RETURN

P3

40A

PO3

P4

40A

POWER_RETURN

P5

40A

POWER_RETURN

LP2

20A

PO2

P6

40A

PO1

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Power & Energy Efficiency

Module Pitch (height)

Max Secondary Side

Max Primary Side

Comment

1.0

.310

.660

Reference

1.1

.360

.710

AFB,CC

1.2(a)

.360

.810

AFB,CC

1.2(b)

.510

.660

AFB,CC

1.3

.610

.660

AC,CC

1.48

.310

1.160

AFT

1.6

.560

1.010

CC

1.7

.51

1.160

CC

The most recent release of VITA 62 adds a number of different pitches (heights). This addition is especially important when incorporating hold-up, as needed components might require additional height.

The process of tailoring a system can bring many desirable features. Some of those features include special over-voltage, over-current, temperature warnings and limits. It is quite common to see wide variations in these requirements from system to system.

And it would be quite difficult and expensive to have each variation fixed in hardware. The ability to program these settings lets standard hardware handle such modifications.

Temperature warnings and limits are other factors that are often system dependent. The ability to read output and input currents can also help in monitoring supply operation and in diagnosing system problems.

VITA 62 is an important step forward in the effort to have an open VPX standard power supply. But there is still significant work necessary to meet new and existing system needs. Established standards don’t cover several hardware areas well, and software definitions need to be addressed.

In hardware, there is no real definition of an energy storage card that would let system designers and manufacturers have a standard configuration.

While VITA 62 does mention these devices, there is no connector pin assignment. As a minimum, such a card

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CUSTOMIZED APPROACH

needs definitions for input and output pins. It would also be desirable to have basic logic and control functions defined, such as “card ready” or “fault.”

Another need is a specification for three-phase inputs for 3U cards. The current 3U connector lacks enough input pins for this function. The original VITA 62 committee did not foresee a need for these inputs, but many systems now have such a requirement. There is, however, an effort underway to tie up some of these loose ends.

Industry is also paying more attention to the VPX arena.

The DOD “Host” for hardware and “Fast” for software are just two of the industry thrusts to reinforce the Open VPX Standard. The most recent release of VITA 62 adds a number of different pitches (heights). This change is especially important when incorporating hold-up, as the required components might need additional space.

Behlman actively participates in VITA committees and offers a wide variety of both standard and existing nonstandard VPX power cards, while also giving system designers the opportunity to reconfigure cards for maximum performance and economy. For example, working within its new VPXtra Reconfiguration Program, Behlman has modified the VPXtra 500M to create the new VPXtra500M1. It has 80 A of 5 Vdc output: 40 A on the designated PO3 pin and the additional 40 A on the PO1 pin, in place of the 12 Vdc. This unit, in conjunction with the standard Patra 500M, can supply 120 A of 5 Vdc along with the customer-required 12 Vdc and 3.3 Vdc.

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Power & Energy Efficiency

Gauging energy efficiency in Complex Motor Drives KEN JOHNSON TELEDYNE LECROY

Modern instrumentation can help reveal sources of power dissipation problems where electric motors are tightly integrated with their controls and test points are hard to find.

LARGE

integral-horsepower motors account for about 90% of the electricity consumed by all motors. But large motors make up only 10% of all motor sales. The other 90% are fractional horsepower motors as found in power tools, industrial automation, elevators, and vehicles. Many of these smaller motors are characterized by a tight integration of the control system, motor drive, and motor itself. This integration forces engineers to take similarly integrated measurements of the control system, motor drive, and motor when debugging and validating the system.

It can be useful to see how modern instrumentation provides facilities for making these integrated measurements and for sniffing out energy efficiency problems. One instrumentation system designed for making these kinds of measurements is the Teledyne LeCroy MDA810 Motor Drive Analyzer. The following measurement examples demonstrate how one might use this analyzer to test a small hand-held tool containing a sine-modulated permanent-magnet synchronous motor operating at high speed.

Visible in this display from the MDA810 is the control signal for rotational direction reversal as well as various calculated numerics and statistics values. The acquired signals are the voltages C1 and C2, currents C5 and C6, and control C4. In this example, the values of greatest interest are the motor VRMS, IRMS, real power, apparent power, reactive power, power factor, and phase angle. The mean values of these measurement parameters for the full acquisition appear in the Numerics table, similar to what a power analyzer would display. The P(∑rst ) and S(∑rst) waveforms (overlaid on top of each other in the bottom right grid) are per-cycle “synthesized” waveforms that plot the per-cycle values (shown in the Statistics table) versus time. These percycle values are time-correlated to the original acquisition waveforms. They are created by touching or clicking a Numerics table cell value.

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GAUGING ENERGY EFFICIENCY The tool’s operation involves reversal of the motor direction once per second. The measurements examine how drive signals correlate to the behavior of the tool and dynamic power consumption. Measurements include the power consumption during the transition in rotational direction and the difference in power consumption while rotating in each direction. The goal: to understand and mitigate power losses during these periods, which, if too high, could annoy users and/or cause reliability issues.

A typical measurement scenario might use five 12-bit, 1-GHz analog acquisition channels. They would monitor two control signals, the encoder position of the rotor, the actual velocity of the motor, and the commanded velocity. An external controller board designed by the test engineer can process the encoder position and velocity signals for input to the MDA810.

Because this is a sensorless motor, the encoder is added for test purposes only – it’s not part of the motor in normal operation.

The control signals on channels 1 and 2 tell the motor to change rotational direction. The rising edge of channel 1 initiates the reversal in motor rotation, while

OUTPUT POWER ANALYSIS In this example, we use a two-wattmeter method to analyze the motor drive output and calculate threephase power values. The two-wattmeter method permits measurement of three-phase system power with only four signals, leaving other channels available to acquire control or power behaviors. The MDA810 also supports three-wattmeter methods.

In the two-wattmeter method, two high-voltage differential probes and two current probes connect to the motor drive output. A single acquisition of the lineto-line voltage and line current waveforms would show both to be 120° out of phase, normal in a three-phase system. While the un-zoomed waveform appears to be noisy, a zoomed view reveals that the appearance of noise is really the switching characteristics of the devices in the drive output. These details could not be seen with a traditional eight-bit oscilloscope, but the MDA810’s 12-bit acquisition system has enough resolution to make this observation possible.

A longer acquisition captures the complete motor rotational direction change and enables calculation of

In this screen from the MDA810, Zoom+Gate is set to display one motor rotational operating cycle. The ∑abc parameters in the Numerics table represent the full-spectrum power values, while the ∑rst parameters represent the fundamental only.

Using cursors, the operator can measure the time of the motor operating cycle and then calculate dissipated energy from the power parameters. The power value shown in the Numerics table for ∑abc is 3.668 W. The length of time measured using cursors is 907.68 msec. This value is represented as the ΔX value located in the bottom right corner of the screen.

the falling edge of channel 2 represents the point in time when the motor completes its reversal. Capturing data over a longer period (five seconds, in this example) enables the viewing of many cycles of the transition.

Zoom traces display details of one of these transitions, clearly showing the timing of the control signals and the motor’s response. We also monitor speed change at this transition point. From these signals, we see that the motor rotational reversal behaves well and operates as expected.

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power values before, during, and after the change. Of interest is the amount of power consumed during the transition from one direction to another. Ideally, we do not want a sharp increase in power at this transition point. This longer acquisition contains two transitions of the motor direction.

To determine the cyclic period for making all voltage, current, and power calculations, we must choose a signal to contain the “reference period.” In the MDA810, we refer to this as the “sync” signal. The sync

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Power & Energy Efficiency

determines the measurement interval for computing the per-cycle voltage, current, power, efficiency, mechanical measurements, and other values. It is usually necessary to filter the sync signal to remove high-frequency content to get better periodicity, and this task is simple in the MDA810.

One may use the line-to-line voltage, filtered with a 500-Hz lowpass filter, as the sync signal. We examine the sync signal to verify it identifies the right time periods, thereby ensuring power calculations are correct. After verifying the measurement periods, we turn off the sync signal.

In this example, the values of greatest interest are the motor VRMS, IRMS, real power, apparent power, reactive power, power factor, and phase angle. The mean values of these measurement parameters for the full acquisition appear in a Numerics table display in the accompanying figure, similar to what a power analyzer would display. The P(∑rst) and S(∑rst) waveforms (overlaid on top of each other in the bottom right grid) are per-cycle “synthesized” waveforms that plot the per-cycle values (shown in the Statistics table)

versus time. These per-cycle values are time-correlated to the original acquisition waveforms. They are created by touching or clicking a Numerics table cell value.

These per-cycle waveforms clearly indicate the dynamic power behaviors of the motor drive output and motor, something that would not be obvious by only viewing the mean value in a Numerics table.

Viewing the real power and apparent power per-cycle waveforms at the directional transitions of the motor provides valuable insight into the power consumption during each direction change, important in this application because this motor is part of a hand-held tool. Minimizing power consumption keeps the tool from becoming too hot to hold.

For a closer look at the area of interest during one of the transitions, we use the MDA810 Zoom+Gate feature. It provides a simple means of simultaneously zooming all input sources, detailed waveforms, and sync signals. It can also position the zoom window on any portion of the trace. The common zoom window then acts as a measurement gate for the Numerics and Statistics tables.

We focus Zoom+Gate on a given area of interest, such as the transition from one rotational direction to the other. In the example, this is one complete cyclic period as shown by the DrvOutSyncZ sync signal.

The Statistics table shows a power consumption of 3.894 W during the transition. This value is reasonable for the motor under test.

Determination of heat loss during operation provides a still deeper analysis of motor power consumption.

We analyze the loss via the MDA810 by measuring power values using the Harmonic Filter settings at both Full-Spectrum and Fundamental simultaneously. After comparing these results, we calculate the heat loss in the windings as the difference between the Full Spectrum and Fundamental calculated real power results. The Harmonic Filter setup defines the filter applied to the input waveforms for power calculations. We may define this filter in both the ac input and drive output.

For this example, the ac input harmonic filter is set for Full Spectrum while the Drive Output is set for Fundamental only. We set the voltage and current inputs to be on the same

This screen from the MDA810 shows motor position (C3: encoder position, C7: sensorless position), control (C4), speed (C8), and power (P(∑rst) and S(∑rst)) signals. Note that in the clockwise rotation, the per-cycle P(∑rst) and S(∑rst) power waveforms are smooth and consistent. During counterclockwise rotation, however, the power waveforms in the upper right corner reveal an oscillation in the signal.

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GAUGING ENERGY EFFICIENCY

channels in both setup dialogs. As a result, the only difference in the power calculations is the measured harmonics.

The accompanying figure shows Zoom+Gate set for one motor rotational operating cycle. The ∑abc parameters in the Numerics table represent the fullspectrum power values, while the ∑rst parameters represent the fundamental only. Using cursors, we can measure the time of the motor operating cycle and then calculate Joules from the power parameters. The equation for converting watts to Joules is: E(J) = P(W) × t(s) For this example, the power value shown in the Numerics table for ∑abc is 3.668 W. The length of time measured using cursors is 907.68 msec. This parameter displays as the ΔX value located in the bottom right corner of the accompanying figure. Converting to Joules, the energy consumed for ∑abc is 3.33 J. Using the same method to find ∑rst, we calculate 2.51 J was consumed.

By subtracting these numbers, we arrive at 0.82 J. The 0.82 J value represents the heat loss in the winding for this tool while it rotates in one direction before switching directions.

Let’s look at an example where the power plots indicate an issue with the motor operation. In this next example, we test the same type of motor with the same two-wattmeter wiring configuration.

The display also shows motor position (C3: encoder position, C7: sensorless position), control (C4), speed (C8), and power (P(∑rst) and S(∑rst)) signals. Note that in the clockwise rotation, the percycle P(∑rst) and S(∑rst) power waveforms are smooth and consistent. When rotating counterclockwise, however, the power waveforms in the upper right corner reveal an oscillation in the signal.

After observing this vibration, further investigation discovered an issue with the motor commutation, which stemmed from a sensorless control issue.

REFERENCESS Teledyne LeCroy www.teledynelecroy.com

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Publisher Mike Emich memich@wtwhmedia.com 508.446.1823 @wtwh_memich Managing Director Scott McCafferty smccafferty@wtwhmedia.com 310.279.3844 @SMMcCafferty EVP Marshall Matheson mmatheson@wtwhmedia.com 805.895.3609 @mmatheson

Courtney Seel cseel@wtwhmedia.com 440.523.1685 @wtwh_CSeel

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MEASURE. EVALUATE. VISUALIZE & CONFIGURE.

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